THE CHINESE UNIVERSITY OF HONG KONG
RFID: Reader Design
Cheung Hing Hung
香港中文大學電子工程學系 DEPARTMENT OF ELECTRONIC ENGINEERING
RFID: Reader Design
Author: Student I.D.: Supervisor: Associate Examiner:
Cheung Hing Hung 02668093 Professor K.K.Cheng Professor K.T.Chan
A project report presented to the Chinese University of Hong Kong in partial fulfillment of the Degree of Bachelor of Engineering
Department of Electronic Engineering The Chinese University of Hong Kong April 2006 2
A. Abstract RFID (Radio Frequency Identification), also called electronic labeling, is a new wireless technology. Reader (Interrogator) can read from or write to a specified identity tag without any mechanic or optical contact whereas it is done by radio frequency communication.
An RFID system can be simplified into three parts; they are data acquisition, reader and tag.
In this project, I am responsible for the RFID interrogator and design the system based on the ISO 18000-6 specification. The interrogator is dedicated to communicate with passive tags.
Interference is a fatal problem in RFID system, therefore, I designed a technique to due with it in order to improve the detecting range, and hence, system performance.
Circuit of individual functional blocks, including VTO, AM modulator, power amplifier, antenna, LNA and mixer, with simulated result and experimental result are evaluated. The data acquisition capability of a simplified transceiver is also tested.
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B. Acknowledgements Professor Cheng Kwok Keung, Michael (B.Sc., Ph.D. (London), MIEEE, AMIEE), who is the supervisor of this project, sets up the goal of this project and gives advice when I faced some technical problems.
Professor Chan Kam Tai (B.Sc. (Hong Kong), Ph.D. (Cornell), MIEEE) is the associate examiner of this project.
Kong Cheuk Pang (M.Phil), who is the tutor of this project, helps me to understand the microwave circuit theory, laboratory instrument, simulation tools and patch antenna theory.
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C. Content A.
Abstract ………………………………………………………………………... P.3
B.
Acknowledgements ……………………………………………………………. P.4
C.
Contents ……………………………………………………………………..… P.5
D.
Introduction………………………………………………………….…………. P.7
E.
Theory and Paper Design
F.
1.
System specification ……………………………………………………… P.9
2.
Estimation of System performance ……………..……………………….... P.13
3.
Challenge and solution ……………………………………………………. P.15
4.
Materials selection …..…………………………………………………..... P.22
Experimental Results 1.
Varactor-Tuned Oscillator ……………..…………..……….……..…......... P.24
2.
AM modulator …………………………………….…..…………………... P.30
3.
Power Amplifier ........................................................................................... P.32
4.
Antenna ……………………………………………….…..………………. P.36
5.
Variable attenuator …………………………...……...................………….. P.40
6.
Voltage Limiter ……………………………………….….………….…….. P.43
7.
LNA ……………………………………………………………………….. P.45
8.
Mixer ………………………….….....………....…………………………... P.49
5
9.
A simplified Interrogator ……….....………....…………………………... P.53
G.
Cost Summary ………………………………………….….….……..…………. P.55
H.
Discussions and Conclusions …………………………..….…..……………….. P.58
I.
Reference ……………………………………………….……...……………….. P.60
J.
Appendices 1.
Equation of maximum detection range…………………………….…...…. P.61
2.
Simulation result of 16-element patch antenna array ……..………….…… P.62
3.
Stabilization of HBFP0405 ………………………………………..…. …... P.65
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D. Introduction The reader is a radio frequency transceiver therefore the design of parameters must fulfill the radio regulation of local government, on the other hand, International Organization of Standard (ISO) has designed a clear standard on RFID system, and the name of the document is ISO 18000-6 [7].
Firstly, one of the specifications is chosen from the document based on the functionality of desired system. Then, the feasibility and performance of the overall system should be verified carefully. After it, the specification of individual functional block set can be confirmed.
Secondly, materials used by the project are selected. The most critical one is apparently substrate, and other components include microwave transistors, diodes, resistors, inductor, capacitors, IC chips, SMT connectors and cable.
Firdly, the circuit design of individual block set is concerned. The circuits are tested under simulation tools like Agilent Advanced Design System (ADS) and Zeland IE3D. ADS is usually used in transmission line and components related circuit whereas IE3D is used in Microstrip Antenna design.
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Fourthly, individually functional block is fabricated and their performance are vertified.
Finally, integrate all the functional blocks into a complete transceiver. The signal transmission, reception and data acquisition capability are tested.
In this project, I have simulated all the functional blocks that are required in my design of RFID transceiver. The functional blocks include VTO, modulator, power amplifier, antenna, LNA, variable attenuator, voltage limiter and mixer. In which, VTO, modulator, power amplifier and mixer are fabricated. A simplified complete transceiver is also built to test the data acquisition capability of my design.
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E. Theory and Paper Design 1.
System specification To begin with, specification of passive backscatter RFID is checked from the ISO
18000-6 document [7]. Table 1.1.1 shows two types of RFID system.
Table 1.1.1
Type Type B can support more tags and have larger potential in logistic industry therefore it is selected. There are many parameters but only Modulation index and Data rate is related to my
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design. They affect the design of modulator and bandwidth of my reader respectively.
Modulation Index 18% modulation index is chosen because it gives a relatively stable power supply to the passive tag.
Data Rate 10 kbps data rate is chosen because another student, who is responsible for data acquisition part, plan to build a slow speed system first. In fact, no matter the selected data rate is 10 or 40 kbps, the turn out bandwidth is still negligible in the radio frequency transceiver section.
Carrier Frequency Another important parameter is the carrier frequency. Base on ISO specification, carrier frequency can be 135 kHz, 13.56 MHz, 433 MHz, 860-960 MHz and 2.45GHz. 860-960MHz is chosen because antenna gain is higher, size of circuit is more compact and antenna is smaller at high frequency, but 2.45GHz is not considered because tolerance and side effect during fabrication can be high for such a high frequency. On the other hand, local radio regulation further restricts the range to 865-868 MHz & 920-925 MHz, 866 MHz is selected
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as the carrier frequency in my design finally.
EIRP By local radio regulation, maximum EIRP is limited by 4W, which is 36dBm. My design will fully meet this limit in order to provide sufficient power to drive the passive tags.
DC Power Supply 9V battery is used so that the reader can be portable.
The interrogator Figure 1.1.1 shows a simplified version of the schematic of the interrogator. Simplified version of the schematic of the interrogator
Figure 1.1.1 11
The transmitting path of the transceiver work as follow: 1. The oscillator gives a power of 10dBm and then it is pre-amplified to 18dBm. 2. Negligible power, about 0dBm, is coupled to the mixer by the non-ideal isolation of a high impedance capacitor. 3. Most of the power is passed to the AM modulator. 4. The power is further amplified to 26dBm by power amplifier. 5. A quadrature coupler is used to imitate an circulator. The isolated port is connected to the receiver whereas the two output ports are used to drive an antenna 6. The antenna, with 10dB gain, gives an EIRP of 36dBm, which is the maximum power limit of a RFID transmitter.
The receiving path of the transceiver work as follow: 1. Received power from the antenna is passed to the quadrature coupler. Half of the power is lost to the transmiting path and half of the power is passed to the receiving path 2. Received signal is amplified by LNA, interference are suppressed and output power is automatically adjusted by AGC before passing to the mixer. 3. Received signal is passed to the mixer and then down-convert to base-band signal directly.
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2. Estimation of System performance Let R be reader, T be tag, d be the distance between the reader and the tag, S be the received signal power, n be the minimum SNR for detection divided by the isolation between the transmitting path and the receiving path. The maximum detection range is ⎛1⎞ Max(d ) = ⎜ ⎟ ⎝n⎠
1 4
×
G R × GT × λ
#1
4×π
(Proof in Appendix 1)
Table 1.2.1 lists the maximum detection range of the interrogator, based on #1 with different inputs. Maximum detection range of the interrogator GR
GT
n
Max (d)
10
1
1.00E-03
0.49022 m
10
1
1.00E-04
0.87175 m
10
1
1.00E-05
1.55022 m
10
1
1.00E-06
2.75673 m
10
1
1.00E-07
4.90223 m
1
0.1
1.00E-03
0.04902 m
Table 1.2.1 The n parameter is critical to the system performance, a value of 10-3 can be achieved easily, but a value of 10-6 or 10-7 is challenging. Details will be discussed in next section. 13
The last row of the table shows a bad case with reader antenna gain of 0 dB, tag antenna gain of -10 dB and n = 10-3, the maximum detection range is about 5 cm only.
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3. Challenge and solution RFID system works like a radar system with amplitude modulation, it greatly simplify the modulation and demodulation processes, especially in the design of tags, it is very important if a cheap and small tag is desired. However, interference becomes a big problem because transmitted signal and returned signal compose of the same carrier frequency; on the other hand, phase is not concerned; therefore, they are only different in amplitude. Signal must has a larger power than interference in order to be received. i.e. S ≥ n × PR where n 0 ∂ω ω =ωO
#7
In many case, dR L (ω ) =0 dω
#8
Then, #7 is simplified to R IN ( A) = − R O (1 −
A ) AM
#9
Power delivered to RL is
P=
1 1 2 Re[VI * ] = I RIN ( A) 2 2
By solving RL =
#10
dP , a convenient value of RL , which maximizes the oscillator power, is dA
RO 3
#11
The theory of a two-port negative-resistance oscillator is very close to that of a one-port negative-resistance oscillator. The procedure is: 1. Firstly, use a potentially unstable transistor at the frequency of oscillation. 2. Design the terminating network to make ΓIN > 1 . Series or shunt feedback can be used to increase ΓIN i.e. make the RF transistor more unstable at the desired frequency 3. Design the load network to resonate Z IN , and to satisfy the start of oscillation condition in (5.2.22). That is
X L (ω O ) = − X IN (ω O )
and
RL =
RO 3
#12 25
In order to make the oscillator tunable, the design of load network is a composite of strip line and a capacitor. The capacitor is replaced by a varactor diode finally so that the output frequency is tunable by varying the property of load network
Figure 2.1.1 is the schematic diagram of the VTO The schematic of the VTO is
Figure 2.1.1
The RF transistor is at the middle. A short circuit stub, at the base of the transistor, imitates an inductor to make the transistor more unstable at the desired frequency. On left hand side, it is the terminating network. On the right hand side, the load network is a composite of strip line and capacitor. The bottom-right capacitor will be replaced by a varactor diode so that the oscillating frequency can be tuned by voltage.
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Figure 2.1.2 shows layout of the VTO The layout of the VTO
Figure 2.1.2
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Figure 2.1.3 shows the relationship between output frequency and tuning voltage. When the tuning voltage increased from 0V to 9V, the output frequency increased from 610MHz to 925MHz. At the desired frequency, 866MHz, the tuning voltage is six volts.
Frequency vs Control-Voltage
Figure 2.1.3
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Figure 2.1.4 shows the output power of the VTO across the tunable spectrum. The output power of fundamental frequency and first harmonic are 9dBm and -13dBm respectively. The power difference between fundamental frequency and first harmonic is 21dB, which is good.
The power of fundamental frequency and first harmonic
Figure 2.1.4
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2. AM Modulator
The AM modulator modulates the carrier with two discrete amplitudes, in fact, it is simply a variable attenuator switched by PIN diodes. Because 18 % modulation index is selected, the two amplitude levels have the ratio of 1: 0.835. Figure 2.2.1 shows the layout of the modulator Layout of AM modulator
Figure 2.2.1
Control voltages are used to turn on or turn of two pairs of PIN diodes, hence, allow RF signal to pass through either the lower path of the upper path. The lower path does not attenuate the signal whereas the upper path gives attenuate the signal with S21=0.835 by a T attenuator.
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Table 2.2.1 shows the measurement of the modulator. The reflection coefficient remains lower than -16dB therefore the circuit is matched to both input and output all the time. The transmission coefficient has a different of -1.48 dB i.e. 0.843 in magnitude, which is quite close to desired value, 1:0.835. Measurement of AM modulator Digital Siganl
High (dB)
Low (dB)
S11
-16.69
-20.4
S21
-0.96
-2.44
S12
-0.99
-2.48
S22
-16.57
-20.55
Table 2.2.1
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3. Power Amplifier
For the design of power amplifier, the choice the transistor is very critical. Unfortunately, there are limited choices from the market. The RF transistor being used is BFQ-19S. It has a maximum power dissipation of 1W whereas my target output power is 400mW. The bias condition is different from other small signal application. Large collector-emitter voltage and collector current is selected in order to deliver high power. In my design, Vce of 8V and Ic of 70mA is used. In the traditional design of power amplifier, the source and load matching network is selected to have low impedance values so that the power amplifier gives high output power provided that voltage or current is fixed. However, since I am going to build a complete transceiver, an un-matched functional block will affect the properties of other functional blocks and complicate the whole circuit design, therefore, I chose to match both input and output port to the transmission line. Fortunately, the performance of the power amplifier does not deteriorate much.
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Figure 2.3.1 shows the layout of the power amplifier Layout of power amplifier
Figure 2.3.1
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Figure 2.3.2 shows the S-parameters of the power amplifier. The reflection coefficient of input and output ports are -17.6dB and -16.44dB respectively, therefore they are matched.
S-parameters of power amplifier
Figure 2.3.2
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Figure 2.3.3 shows the relationship between input power and output power. The power amplifier has a maximum output power of 24.3dBm and a gain of 9dB. Output power vs Input power
Figure 2.3.3
The power dissipation of the power amplifier = Vce × I = 7.5 × 0.1 = 750mW The maximum RF output power of fundamental frequency = 24.3dBm = 270mW The efficiency of the power amplifier = 270 / 750 = 36%
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4. Antenna
As discussed in Materials Selection section, substrate with 3.2 mm thickness and an antenna of 10 dB gain is required by the system specification. Performance of signal patch antenna is not satisfactory therefore patch array is used. Figure 2.4.1 shows the E-field, voltage, current and impedance distribution inside a rectangular patch antenna. Properties of single patch antenna
Figure 2.4.1
Length is λ /2 for the desired frequency to resonate. W is optimized to #13 in order to maximize the gain [4]. W= 2 fo
c (ε r + 1) 2
#13
Most current flow in the middle but not along the edge, therefore an impedance wall is formed at the edge; proper impedance matching is required in order to feed the patch. 36
Figure 2.4.2 shows a linear 4-element array, Linear array of four elements
Figure 2.4.2
Length of both patch and feed line are λ /2 so that all patches resonate in phase. From right to left, the impedance of the feed line is transformed from a high value to a low value, hence, lower radiating resistance and higher radiating power.
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Figure 2.4.3 shows a 16-element antenna array; all feed lines are proper designed so that all rectangular patches resonate in phase. Patch array of 16 elements
Figure 2.4.3
Simulating result of 16 elements patch array antenna is shown in appendix 2
Table 2.4.1 shows the simulated result of various number of arrays. The efficiency is limited by the quality of the substrate and the relatively low carrier frequency. The size of the antenna array can be greatly reduced by using a higher carrier frequency, say 2.4Ghz, whereas
38
the theory of design is almost the same. The gain of the 16 patches antenna is 1.5 dB above the 10 dB target. The excess gain can be reserved to compensate other loss in the circuit Simulating result with different number of elements
number of patches
Gain
efficiency
size
1
~ 1.5 dB
~ 33 %
~ 11 * 9 cm
4
~ 6.5 dB
~ 32 %
~ 11 * 65 cm
16
~ 11.5 dB
~ 31 %
~ 60 * 65 cm
Table 2.4.1
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5.
Variable Attenuator
Bridged-T attenuator [6] is used as the basis of attenuator because only two variable resistors are required and it is achieved by replacing variable resistors by PIN diodes. Figure 2.5.1 shows the circuit of a bridged-T attenuator and a voltage-controlled variable attenuator Bridged-T attenuator
Figure 2.5.1
Where
R1 = Z O (10
L 20
− 1)
and
ZO
R4 = 10
L 20
#14
−1
40
In my design, three bridged-T attenuators are cascaded in order to enhance the performance. The schematic diagram is shown in Figure 2.5.2 Schematic diagram of variable attenuator
Figure 2.5.2
Figure 2.5.3 shows the relationship between transmission coefficient and control-voltage. Attenuation changes with control voltage. From 0V to 8V, the attenuation varies from about 20 dB to 80 dB, therefore a dynamic range of 60 dB. The dynamic range implies the variation of incoming signal that the system can handle. Signals, that out of the dynamic range, will either saturate the amplifier or not strong enough to be detected. The minimum attenuator is about 20 dB, it sounds awful. In fact, it will not downgrade the signal to noise ratio because white noise is negligible. However, a more powerful LNA is required to boost the power again.
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S21 vs control-voltage
Figure 2.5.3
Figure 2.5.4 shows the transient response with different control-voltage. The phase remains constant with different attenuations, it is very important to ensure the 2nπ + π interference suppression loop works. Transient analysis vs control-voltage
Figure 2.5.4 42
6.
Voltage Limiter
This device works like a DC voltage regulator; it gives a constant AC output regardless of the power of input signal. Figure 2.6.1 shows the schematic diagram of a voltage limiter. Schematic diagram of the voltage limiter
Figure 2.6.1
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Figure 2.6.2 shows the simulated result of the voltage limiter. The upper graph is the input voltage with different magnitude whereas the lower graph is the output. Power is grounded by PIN diode if the input voltage is either too positive or too negative; therefore, the output remains almost constant regardless of input power. Simulating result of voltage limiter
Figure 2.6.2
Besides, the output voltage amplitude depends on the forward threshold voltage of the PIN diodes. Again, the output phase remains the same, it is very important to ensure the 2nπ + π loop works.
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7. Low Noise Amplifier (LNA)
As discussed in Challenge and Solution, white noise is negligible in the system design, therefore LNA is simply a high gain amplifier. In this state, the actual gain we required is not confirmed because it depends on the isolation of interference from LNA and the minimum attenuation of variable attenuator, however, a high gain amplifier is usually helpful. In order to obtain high gain and high output voltage swing, two transistors are used. The first one is HBFP0405, which has very high gain but low output voltage; the second one is BFR183, which has relatively low gain but high output power. Table 2.7.1 shows the bias condition of the two RF transistor. Bias condition of HBFP0405 and BFR183
HBFP0405 BFR183 Vce
2V
4.5 V
Ic
5 mA
15 mA
Table 2.7.1
Unfortunately, HBFP0405 is unstable from 0 – 7.3 GHz. In order to prevent it from oscillating, a 250Ω shunt resistor is added to the output to stabilize the transistor. The resistor seldom put at the input of the transistor because it will make the output very noisy.
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The corresponding stability diagram is shown in Appendix 3 and the schematic diagram of a stabilized HBFP0405 transistor is shown in Figure 2.7.1 Schematic diagram of a stabilized HBFP0405
Figure 2.7.1
Table 2.7.2 shows the S-parameters of the two RF transistors after proper biasing and stabilization. S parameter of the two transistors
polar
HBFP0405
BFR183
S11
0.779 / -40.199
0.091 / -178.41
S12
0.020 / 69.763
0.118 / 72.115
S21
9.569 / 155.77
4.507 / 80.900
S22
0.622 / -13.571
0.391 / -24.389
Table 2.7.2
For HBFP0405, S12 is very low, therefore, unilateral conjugate matching is used, i.e. ΓS = S11* = 0.779∠40.199 * ΓL = S 22 = 0.622∠13.571
#15 46
GTU ,max =
1 1 − S11
2
S 21
1
2
1 − S 22
2
= 379.86 = 25.8dB
#16
For BFR183, S12 is quite large; therefore, bilateral conjugate matching is used,
Γ MS = Γ ML =
B1 ±
B 12 − 4 C 1 2C 1
B2 ±
B 22 − 4 C 2
#17
2
2C 2
B1 = 1 + S11 − S 22 − ∆
2
B2 = 1 + S 22 − S11 − ∆ * C1 = S11 − ∆S 22 C 2 = S 22 − ∆S11*
2
2
2
2
Where
2
2
#18
And ∆ = S11 S 22 − S12 S 21 1 − S11 − S 22 + ∆ 2
K=
2
#19 2
#20
2 S12 S 21 ΓMS = 0.691∠181
Solving #6, #7, #8 and #9, finally gives
And GT ,max =
S 21 S12
ΓML = 0.783∠24.9 K = 1.02
( K − K 2 − 1) = 31.28 = 15dB
Cascading the two amplifiers, the total gain = 25.8 + 15 = 40.8 dB
#21
#22
#23
Making use of the four Γ values obtained, all ports of the two transistors are matched by microstrip. Figure 2.7.2 shows the schematic diagram of two-state LNA
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Schematic diagram of two-state LNA
Figure 2.7.2
Figure 2.7.3 shows the gain of the two-state LNA verse frequency. The gain is 40.262dB at the desired frequency, which is very close to the calculated result in #12 that is 40.8 dB. Gain of LNA verse frequency
Figure 2.7.3
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8. Single-Balanced Mixer
A mixer is a three-port network with two input, local oscillator and radio frequency, and one output, intermediate frequency. Mixers are commonly used to multiply signals of different frequencies in an effort to achieve frequency translation. A mixer can be divided into three sections. A simple block diagram of a general mixer is depicted in Figure 2.8.1 Structure of a general mixer
Figure 2.8.1
The first section combine or multiply LO signal with RF signal, in addition, isolation between LO and RF is very important, otherwise, LO signal will contribute noise to RF signal or RF signal will distort other functional blocks in the system. The second section is a non-linear device, which produce multiple frequency components from the combined signal, LO and RF. Diodes and transistors, with proper biasing, are commonly used as the non-liner device. The last section is an IF filter, on one hand, it extracts the desired frequency component from those frequency components generated by the non-linear device; on the other hand, it 49
suppress unwanted frequency components.
In my design, I chose to use passive single-balanced mixer because, on one hand, it has higher efficiency than single-ended mixer; on the other hand, the double-balanced and active properties is not required because the effect of spurious mode is small and the RF signal is sufficiently strong. Figure 2.8.1 depicts the schematic diagram of the mixer. Schematic diagram of the mixer
Figure 2.8.1
A quadrature coupler acts as the combiner of LO and RF signal. Because RF and LO signal compose of the same frequency, a hybrid coupler is an ideal choice to combine the two signals, in addition, provide excellent isolation between the two input port, LO and RF. A pair of diodes acts as the non-linear device. The products are base-band signal, fundamental frequency and harmonics only because the mixer is a direct conversion mixer. The two-way configuration cancels the 90 O phase different of the output of the quadrature coupler. 50
The last section, IF filter, is simply a low pass filter with a large capacitor. It is easy to extract the IF signal, base-band signal, because the closest unwanted frequency is 866MHz, which is very large compared with the wanted signal, proximately 10kHz.
Figure 2.8.2 depicts the layout of the mixer. Strip-lines of the quadrature coupler are bent in order to reduce the size. Layout of the mixer
Figure 2.8.3
Figure 2.8.4 depicts the power of IF, which is DC, with different input power of LO and RF. For a fixed LO power, 0dBm or 10dBm, the IF output power increases with RF-input power. Nevertheless, if RF power is too low, the IF output stop dropping due to non-ideal
51
isolation between LO and RF; contrary, if the RF power is too high, there is a weird behavior at the IF output because the effect of harmonics emerges. IF (i.e. DC) output power with different LO and RF power
0 -40
-30
-20
-10
0
10
IF, DC (dBm)
-10 -20 -30 -40 -50 -60 RF input power (dBm)
LO=0dBm LO=-10dBm
Figure 2.8.4
The measurement result in Figure 2.8.4 stands on the ground of a DC output. Whatever, it should also work in the case 10kHz base-band signal, the data rate of the RFID interrogator, because 10kHz is in the pass-band of the IF filter, which is a low-pass filter.
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9. A simplified interrogator
In order to test the data acquisition capability of the system, a simplified interrogator is built. The interrogator contains five major functional blocks. The layout is depicted in Figure 2.9.1 A Simplified Interrogator
Figure 2.9.1
The first functional block is a VTO, which gives the carrier signal. Most of the power goes to the transmission path and some power couples to the mixer as the LO signal. The second one is modulator. In order to make the result more obvious, a modulator with 100% modulation index is used. The third one is a power amplifier. 53
The fourth one is the quadrature coupler, which imitate a circulator. The two output ports are terminated with an almost-match load, which little amount of incoming signal is reflected. The weak returned signal imitates the back-scattered signal from tags. The last one is a mixer, which converts the RF signal back to base-band signal. Due to non-ideal isolation of the quadrature coupler and incomplete matching of the two “almost matched” loads, the RF signal goes to the mixer is sufficiently strong, therefore, LNA is not required in the simplified interrogator. Figure 2.9.2 depicts the experimental result of the transceiver Modulating signal and Demodulated signal
Figure 2.9.2
The demodulated signal is weak and has much noise. One of the reasons is, although individual functional block works properly, their measurements stand on the ground that both source and load are 50Ω . However, when they are inter-connected, input and output impedance of one functional block may affect that of other function blocks.
54
G. Cost Summary This project involves many RF and microwave discrete components.
RF transistors and diodes are bought from “www.rshongkong.com”, an electronic products retailer. In fact, most of the components are cheap but buying each of them individually is expensive. These components are about 60% cheaper if the purchase quantity is more than 100. The price can be further reduced for a more bulky purchase.
Other materials, like dielectric substrate, SMT connectors, chip resistors, chip inductors and chip capacitors can be obtained from microwave laboratory or supervisor.
55
Table 3.1.1 lists some of the major materials and components used in this project. Table of expenditure Materials and components
FR4 substrate
Quantities
Cost (HKD)
~ 4 pieces of A3 size
$0 (from laboratory)
SMT connector
5
$0 (from supervisor)
RF transistor, BFR-183
15
$8.21
RF transistor, BFQ-19S
15
$10.20
PIN diode, BAR-64-05
8
$10.20
Tuning diode, BB-833
5
$5.84
Schottky Diode BAT-62
3
$15.70
Chip resistors, inductors and capacitors
~ 200
$0 (from laboratory)
solder
---
$0 (from laboratory)
Others
---
$100
Total cost
$534.05
Table 3.1.1
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Equipments used:
Microwave Network Analyzer
Microwave Signal Generator
Microwave Spectrum Analyzer
Cathode Ray Oscilloscope
DC Power Supply
Software used:
Agilent Advanced Design System 2004
IE3D
57
H. Discussions and Conclusions Interference
In the design of RFID interrogator for passive tags, a more complex design of the interrogator trades off for the simple design of passive tags. For example, on one hand, tags do not require generating carrier signal nor doing frequency translation, they can communicate with the interrogator by back scattering. On the other hand, it is very difficult for the interrogator to distinguish back-scattered signal from various types of interference. In order to suppress the interference, hence, increase the detection range, the isolation between the transmitting path and the receiving path is very important. In my design, I have suggested a solution to due with this problem but how good is it is still a question. There is a difficulty to simulate the system because it involves high carrier frequency and base-band algorithm, therefore only transient analysis can be used but the computation power required for the simulation will be extremely large. An alternative is to assume the phase canceling loop works perfect and to use low frequency signal to represent the envelope of the carrier frequency. I have tested the system by this method in software simulator. It works as expected.
Design Technique
Software simulators play an important role through the whole project. Luckily,
58
simulators usually accurately predict the behavior of the circuit provide that the software model of those components exist.
Measurement
When the RF power is large, say, above 20dBm, measurement becomes more difficult, because the maximum power output of the microwave signal generator is 20dBm only. In addition, RF cables become very lossy at high power. The loss can be as high as 5dB, which is a disaster. Other equipments, like network analyzer and spectrum analyzer, are easy to use and give precious data.
What I have learned
I this project, I have learned many things. It strengthens my practical experience in microwave circuit design. Firstly, I learned how to start a project by following the international standard of a product. Secondly, estimate system performance and design parameters for each functional block set. Thirdly, simulate individual functional block by software simulator. Fourthly, carry out measurements by various instruments. Finally, record design procedures and measurement results. Also, writing project.
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I. Reference [1] Guillermo Gonzalez, Microwave Transistor Amplifiers: Analysis and Design, second edition [2] http://www.daycounter.com/Calculators/Complete-RF-Amplifier-Design-Analysis-Calcula tor.phtml [3] http://ihome.cuhk.edu.hk/~s026680/MWcal.htm [4] Constantine A. Balanis, .Antenna theory: analysis and design, 1938. [5] David M. Pozar. Pozar, David M, Microwave engineering [6] http://www.odyseus.nildram.co.uk/RFMicrowave_Circuits_Files/Attenuator.pdf [7] Information technology automatic identification and data capture techniques — Radio frequency identification for item management air interface — Part 6: Parameters for air interface communications at 860-960 MHz, ISO/IEC FDIS 18000-6:2003(E) [8] K. Kurokawa, “Some Basic Characteristics of Broadband Negative Resistance Oscillator Circuits,” The Bell System Technical Journal, July 1969.
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J. Appendices 1. Equation of maximum detection range
Let R be reader, T be tag, d be the distance between reader and tag, S be the received signal power, n be the ratio of transmitting power coupled to LNA (interference). Also, P be power and G be gain. In forward transmission path,
Power density at Tag ( Pd ) =
PR × G R 4×π × d 2
Power received by Tag ( PT ) = Pd × AT = Pd ×
#A GT × λ2 4×π
#B
In backscattering path
Assume the tag backscatter all the power. Power density at Reader ( P0 ) =
PT × GT 4×π × d 2
Power received by Reader ( S ) = P0 × AR = P0 ×
#C G R × λ2 4×π
#D
Solving #A, #B, #C and #D, gives S=
PR × G R GT × λ2 GT G R × λ2 PR × G R2 × GT2 × λ4 × × × = 4×π 4×π 4×π × d 2 4×π × d 2 (4 × π × d ) 4
⎛P ⎞ d =⎜ R ⎟ ⎝ S ⎠
1 4
×
G R × GT × λ 4×π
#E
#F
S ≥ n × PR is necessary in order to detect the backscattered signal. It is discussed in the
Challenge and Solution section, therefore, ⎛1⎞ Max(d ) = ⎜ ⎟ ⎝n⎠
1 4
×
G R × GT × λ 4×π
#1
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2. Simulating result of 16 elements patch array antenna
Figure 4.2.1 shows the return loss, S11, of the 16-element patch antenna array. At the desired frequency, the return loss is 17dB. Return loss (S11)
Figure 4.2.1
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Figure 4.2.2 shows the 3D diagram of the directivity of the 16-element patch antenna array. The directivity is 16.6dBi Directivity
Figure 4.2.2
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Figure 4.3.3 shows the 3D diagram of the gain directivity of the 16-element patch antenna array. The maximum gain is 11.5dBi Gain
Figure 4.3.3
Efficiency of the antenna = 11.52 dB - 16.6 dB = -5.08 dB = 31 %
#J
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3. Stabilization of HBFP0405
Figure 4.3.1 shows the source and load stability circles on smith chart before stabilization. Red circles show the source and load stability circles from 100 MHz to 5 GHz. Circles in other colors show the gain of source and load conjugate matching. They overlap with stability circles. Source and Load stability circles before stabilization
Figure 4.3.1
The amplifier is conditionally stable and unilateral conjugate matching cannot be applied.
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Figure 3.3.2 shows the source and load stability circles on smith chart after stabilization. The amplifier is still conditionally stable, fortunately, unilateral conjugate matching of source and load at 866 MHz are located in the stable region for all the frequencies. Source and Load stability circles after stabilization
Figure 4.3.2
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