FULLY CURRENT CONTROLLABLE AM~slash ... - Semantic Scholar
➠
➡ FULLY CURRENT CONTROLLABLE AM/FM MODULATOR AND QUADRATURE SINUSOIDAL OSCILLATOR BASED ON CCCIIs Montree Siripruchyanun*, Poolsak Koseeyaporn*, Jeerasuda Koseeyaporn**, Paramote Wardkein ** * Department of Teacher Training in Electrical Engineering, Faculty of Technical Education, King Mongkut’s Institute of Technology North Bangkok, Bangsue, Bangkok, 10800, THAILAND Email: [email protected] ** Department of Telecommunication Engineering, Faculty of Engineering, King Mongkut’s Institute of Technology Ladkrabang, Ladkrabang, Bangkok, 10520, THAILAND ABSTRACT In this article, a sinusoidal oscillator, AM and FM signal generator based on translinear current conveyors is introduced. The frequency and amplitude of the proposed circuit can be controlled by the bias currents. When an input current is applied as an information signal to the first and the second CCCII+s (Current Controlled Current Conveyors), the network function as a FM signal generator. Contrarily, an AM signal is obtained by employing such information signal applied to the third CCCII+. In addition, this network simultaneously produces two signals that are 90D different in phase resulting the quadrature sinusoidal signals. This circuit consists of three CCCII+s and two grounded capacitors. Where without any external resistors, this circuit is then suitable for IC architecture. The PSPICE simulation results are depicted. The given results agree well with the theoretical anticipation where the power consumption is approximately 2.4mW. 1. INTRODUCTION A sinusoidal signal is widely used in many areas of measurement, instrumentation, signal processing including electronic and communication systems. In previous literatures of the sinusoidal oscillator, from our investigation, mostly the main objectives are to develop the circuit to obtain; simple circuit description, good stability, high frequency response and frequency adjustable [1-2]. Conversely, amplitude control of sinusoidal oscillator is rarely exhibited. Some proposed sinusoidal oscillators with amplitude control feature [3-4] traditionally use analog multipliers, causing the complicated circuit. Additionally, distortion due to non-linear characteristic of the multipliers can not be avoidable. The initial conditionsrestoration is an alternative method [5] to control amplitude of sinusoidal signal. With this scheme, however, not only the circuit is complicated but also the output signal inevitably encounters with the high harmonic distortion. Recently, the current mode sinusoidal oscillators [6-7] have the attractive features relative to those of the voltage mode such as electronically tunable, low power consumption and high frequency range. Unfortunately, the amplitude of those sinusoidal oscillators can not be arbitrarily controlled since the amplitude is dependent on the oscillated frequency. In this paper, a new current mode sinusoidal oscillator with a new technique to control amplitude is presented. The proposed circuit is based on translinear current controlled conveyors, consisting of 3 positive second generation current controlled
0-7803-8251-X/04/$17.00 ©2004 IEEE
conveyors (CCCII+) and 2 grounded capacitors (benificial to an IC implementation [8]). The outstanding features of the proposed circuit are that; it can simultaneously provide quadrature sinusoidal signals in both voltage and current modes, which is very profitable in communication systems; the oscillation frequency can be controlled by the input bias currents; and the quadrature amplitudes can be linearly controlled by another input bias current whereas no effect is on the condition of oscillation [9-10]. With the proposed principle, it can generate AM and FM signal without an additional device requirement. The PSPICE simulation results are also shown, which are in correspondence with the theoretical analysis. 2. CIRCUIT PRINCIPLE 2.1. The second-generation current controlled conveyor Since the proposed circuit is based on CCCIIs, a briefly review of CCCII is given in this section. Basically, the CCCII is composed of translinear elements, mixed loops and complementary current mirrors [7]. Generally, its properties are similar to the conventional CCII, except that the CCCII has a finite input resistance RX at the X terminal. This intrinsic resistance can be controlled by the bias current I B as shown in the following equation VT
=
RX
(1)
2I B
IB IX
X Z
IZ
Y (a)
X
IB
IX
IZ Z
RX
1
IX
Y (b) Fig. 1 Second generation current controlled conveyor (a) Symbol (b) Equivalent circuit
IV - 549
ISCAS 2004
➡
➡ is the thermal voltage. The symbol and the
From eqn. (10), the response of vo 3 (t ) can be decomposed into
equivalent circuit of the CCCII+ are illustrated in Fig. 1(a) and (b), respectively.
2 parts which are natural response: vo 3n (t ) and forced response:
2.2. The proposed circuit and operation Fig. 2 demonstrates the circuit scheme of quadrature oscillator. The depicted bias currents I B 1 , I B 2 and I B 3 are respectively
vo 3 (t ) = vo 3n (t ) + vo 3 f (t )
Where VT
the input bias currents of CCCII 1, CCCII 2 and CCCII 3. From the circuit and the CCCII properties in section 2.1, The following results are obtained Io1 (t ) = Io 3 (t ) Io2 (t ) = Io 3 (t ) =
(3)
RX 1 vo 2 (t ) RX 2 d v (t ) dt o2
I B1
v (t )¯ ¡¢ o 3 °±
C1
1
(7)
Y
X
I o 2 (t ) C2
2
Z
(14)
vo 3 (t ) = A cos Xot + B sin Xot
(16)
A and B coefficients can be determined by using the initial condition: vo 3 (0) = VSAT . From eqn. (16), we receive
X vo 3 (t )
CCCII +
2I 1 = B RXC VTC
The forced response can be subsequently calculated from eqn. (10) as follows vo 3 f (t ) = 0 (15)
I o 3 (t )
CCCII +
A = VSAT and B = 0
X Z
(13)
Consequently, the total output response; vo 3 (t ) can be shown as
I B2
Z
CCCII +
vo 3n (t ) = A cos Xot + B sin Xot
Xo =
(6)
vo 2 (t )
Y I o1 (t )
(12)
Then,
(5)
d2 vo 3 (t ) = RX 1RX 3C 1C 2 2 vo 2 (t ) dt vo 1 (t )
d2 ¯ ¡v (t )° + vo 3 (t ) = 0 dt 2 ¢ o 3 ±
RX2 C 2
While Xo is oscillation frequency equal to 1/ RXC or
vo1 (t ) = RX 1C 2 RX 3
is a
(4)
and
vo2 (t ) =
Whereas the initial oscillation: vo 3 (0) = VSAT , VSAT
(11)
saturation voltage of the CCCII dependent on power supply voltage. The natural response can be obtained by employing eqn. (12)
(2)
vo1 (t )
RX 2
vo 3 f (t ) as follows [9-10]
(17)
Thus, the completed output responses are
3 Y
IB3
vo 3 (t ) = (VSAT ) cos Xot
(18)
IB 3
(19)
vo2 (t ) =
IB
(VSAT ) cos Xot
Fig. 2 Block diagram of the proposed circuit
Similarly, substituting eqn. (19) into eqn. (5), then it yields By substituting eqn. (6) into (7), it can be shown that d2 vo 3 (t ) = RX 1RX 2C 1C 2 2 ¡¢vo 3 (t )i ¯°± dt
(8)
Thus, RX 1RX 2C 1C 2
d2 ¯ ¡v (t )° + vo 3 (t ) = 0 dt 2 ¢ o 3 ±
vo1 (t ) =
(VSAT ) cos (Xot + 90°)
(20)
Io2 (t ) can be obtained by using eqn. (2)-(4) as follows
as C and I B , respectively). The eqn. (9) becomes d2 ¯ ¡v (t )° + vo 3 (t ) = 0 dt 2 ¢ o 3 ±
IB
The quadrature sinusoidal signals in current mode: Io1 (t ) and (9)
If C 1 and C 2 , I B 1 and I B 2 are forced to be identical (designed
RX2 C 2
IB3
(10)
IV - 550
Io1 (t ) =
Io2 (t ) =
2I B 3 VT 2I B 3 VT
(VSAT ) sin (Xot 90°)
(21)
(VSAT ) cos (Xot + 90°)
(22)
➡
➡ 107
I B 3 is constant and I B is used as a modulating signal, the
106
Similarly, if I B is constant whereas I B 3 is a modulating signal, the quadrature AM signals can be accomplished at either Io1 (t ) and Io2 (t ) or vo1 (t ) and vo2 (t ) . 3. SIMULATED AND EXPERIENTAL RESULTS To prove the performances of the proposed circuit, the PSPICE simulation program was used for the examination. The PNP and NPN transistors employed in the proposed circuit were simulated by respectively using the parameters of the PR200N and NR200N bipolar transistors of ALA400 transistor array from AT&T [11]. Fig. 3 depicts schematic description of the CCCII+ used in the simulations. The power supply voltage was set at ±5V and the capacitors C 1 and C 2 are 10nF. Firstly, the
simulation results are in accordance with the theoretical analysis shown in eqn. (14). VCC Q8 Q10 Q7 Q9
104 103
101 10-7
10-6
10-5
10-4
10-3
Input Bias Currents (A) Fig. 5 Oscillation frequencies against bias currents of CCCIIs in the proposed circuit 8x10-3 Amplitude of current: Io2(t), (A)
depicts the plots of the simulated and theoretical oscillation frequency versus the bias currents: I B 1 , I B 2 . It is seen that the
105
102
input bias currents: I B 1 , I B 2 and I B 3 are respectively set to 150µA , 150µA and 5µA where are shown in Fig. 4. Fig. 5
C=10nF Theory C = 1nF Theory
800x10-3 Theory of Io2 amplitude
7x10-3 6x10
Simulated Amplitude of Io2(t) Theory af Vo2 amplitude Simulated Amplitude of Vo2(t)
-3
5x10-3
600x10-3
4x10-3
400x10-3
3x10-3 2x10-3 1x10
200x10-3
-3
0
Q1 Y
2
4
6
8
10
12
14
16
18
20
Input bias current: IB3 (PA)
Z
X Q3
IB
0 0
Q2
Amplitude of voltage: Vo2(t), (V)
quadrature FM signals can be obtained at Io1 (t ) and Io2 (t ) .
Frequency (Hz)
From eqns. (19) through (22), they should be noted that the amplitude of sinusoidal signal in current-mode and voltagemode can be proportionally controlled by I B 3 . In addition, if
Q4
Fig. 6 Amplitudes of v02 (t ) and I 02 (t ) against bias current: I B 3
Q13
Q13
Q5
Q6 VEE
Fig. 3 Schematic description of the CCCII+
Total Harmonic Distortion(THD)
.08
Q12
.06
.04
.02
0.00
0.0
.2
.4
.6
.8
1.0
Frequency(MHz)
Fig. 7 The simulated result of THD with 10nF capacitors
The simulated results of amplitude control by adjusting I B 3 are Fig. 4 The simulation results of waveforms in the circuit during steady stage Top traces: , Io1 (t ), Io 2 (t ) Bottom traces: , vo1 (t ),
vo 2 (t )
shown in Fig.6. In this case, the input bias currents: I B 1 and I B 2 were set to be 200µA . As predicted from eqns. (19)-(22), the simulated results are in correspondence with the expectation. Fig. 7 insists that the simulated result of THD (Total Harmonic Distortion) is less than 7% with 10nF capacitors. The simulated
IV - 551
➡
➠ results of the proposed circuit serving as an FM and an AM signal generator are also illustrated in Fig. 8 and Fig. 9, respectively. They exhibit a good agreement between the theoretical analysis as shown in eqns. (19)-(22) and the simulation. In addition, to insist that the proposed circuit can operate practically, an experimental result of the proposed circuit employing CA3096s transistor arrays as discrete elements is also depicted in Fig. 10.
4. CONCLUSIONS The quadrature sinusoidal oscillator with amplitude control has been presented. The quadrature signals can be utilized in either current mode or voltage mode. The oscillated frequency can be tuned by adjusting I B 1 and I B 2 over a wide frequency range. Furthermore, the amplitude control can also be achieved by tuning another input bias current: I B 3 . The proposed circuit can generate either quadrature AM or FM signals without an additional element requirement. All of simulation experimental results confirm the theoretical analysis. Since the proposed circuit consists of merely three CCCIIs and two grounded capacitors, it is suitable for monolithic implementation in IC technology. The proposed circuit is expected to be useful for applications in communication, instrumentation and measurement systems, especially at a wide range of frequencies.
5. REFERENCES [1]
Fig. 8 The result of operation as FM signal generator where I B 3 = 5µA;C = 5nF
Fig. 9 The result of operation as AM signal generator where I B 1 = I B 2 = 50µA;C = 5nF
Liu S. I., and Tsay J. H.. “Single-resistance-controlled sinusoidal oscillator using current-feeback amplifiers”. Int. J. Electronics, 80:661-664, 1996. [2] Cicekoglu O., Cam U., and Kuntman H. “Single-resistancecontrolled sinusoidal oscillators employing single FTFN and grounded capacitors”, Proceedings of the 44th IEEE 2001 Midwest Symp. on Circuits and Systems, Dayton, OH, pages 874 –877. August 2001. [3] Filanovsky I. M., Qin S. S., and Kothapalli G. “Sinusoidal oscillator with voltage controlled frequency and amplitude”. Int. J. Electronics, 68:95-112, 1990. [4] Senani R., and Bhaskar D. R. “New active-R sinusoidal VCOs with linear tuning laws”. Int. J. Electronics, 80:57-61, 1996. [5] Pookaiyaudom S., and Saivichit K. “RC phase-shifter variable sinusoidal oscillators using initial conditions-restoration amplitude control”. IEEE Trans. on Instru & Meas., 39:10381044, 1990. [6] Kiranon W., Kesorn J., Sangpisit W., and Kamprasert N. “Electronically tunable multifunctional translinear-C filter and oscillator”. Electronics letters, 33:573-574, 1997. [7] Kiranon W., Kesorn J., and Wardkein P. “Current controlled oscillator based on translinear conveyors”. Electronics letters, 32:1330-1331, 1996. [8] Sun Y., and Fidler J.K. “Synthesis and performance analysis of universal minimum component integrator-based IFLF OTAgrounded capacitor filter”. IEE Proc. Circuit Devices Syst., 143:107-114, 1996. [9] Wardkein P., Wongsuwan C., and Maneechugate T. “The simple sinusoidal oscillator amplitude control by force response”. (in Thai), The 23rd EECON conference, Chaingmai, THAILAND, November 2000, pages 645-648. [10] Tammawut B., Wardkein P., and Maneechugate T. “The simple amplitude control by force response for current mode oscillator”. (in Thai), The 24th EECON conference, Bangkok, THAILAND, November 2001, pages 1018-1023. [11] Frey D. R. “Log-domain filtering: An approach to current-mode filtering”, IEE Proc. G., Circuits Devices Syst., 140:406-416, Dec. 1993.
! !
1) C h 1: 2) C h 2:
100 mVolt 5 us 100 mVolt 5 us
Fig. 10 Experimented result of quadrature output signals in voltage mode where I B 1 = I B 2 = 4.4mA, I B 3 = 0.88µA;C = 0.1µF
IV - 552
➡ FULLY CURRENT CONTROLLABLE AM/FM MODULATOR AND QUADRATURE SINUSOIDAL OSCILLATOR BASED ON CCCIIs Montree Siripruchyanun*, Poolsak Koseeyaporn*, Jeerasuda Koseeyaporn**, Paramote Wardkein ** * Department of Teacher Training in Electrical Engineering, Faculty of Technical Education, King Mongkut’s Institute of Technology North Bangkok, Bangsue, Bangkok, 10800, THAILAND Email: [email protected] ** Department of Telecommunication Engineering, Faculty of Engineering, King Mongkut’s Institute of Technology Ladkrabang, Ladkrabang, Bangkok, 10520, THAILAND ABSTRACT In this article, a sinusoidal oscillator, AM and FM signal generator based on translinear current conveyors is introduced. The frequency and amplitude of the proposed circuit can be controlled by the bias currents. When an input current is applied as an information signal to the first and the second CCCII+s (Current Controlled Current Conveyors), the network function as a FM signal generator. Contrarily, an AM signal is obtained by employing such information signal applied to the third CCCII+. In addition, this network simultaneously produces two signals that are 90D different in phase resulting the quadrature sinusoidal signals. This circuit consists of three CCCII+s and two grounded capacitors. Where without any external resistors, this circuit is then suitable for IC architecture. The PSPICE simulation results are depicted. The given results agree well with the theoretical anticipation where the power consumption is approximately 2.4mW. 1. INTRODUCTION A sinusoidal signal is widely used in many areas of measurement, instrumentation, signal processing including electronic and communication systems. In previous literatures of the sinusoidal oscillator, from our investigation, mostly the main objectives are to develop the circuit to obtain; simple circuit description, good stability, high frequency response and frequency adjustable [1-2]. Conversely, amplitude control of sinusoidal oscillator is rarely exhibited. Some proposed sinusoidal oscillators with amplitude control feature [3-4] traditionally use analog multipliers, causing the complicated circuit. Additionally, distortion due to non-linear characteristic of the multipliers can not be avoidable. The initial conditionsrestoration is an alternative method [5] to control amplitude of sinusoidal signal. With this scheme, however, not only the circuit is complicated but also the output signal inevitably encounters with the high harmonic distortion. Recently, the current mode sinusoidal oscillators [6-7] have the attractive features relative to those of the voltage mode such as electronically tunable, low power consumption and high frequency range. Unfortunately, the amplitude of those sinusoidal oscillators can not be arbitrarily controlled since the amplitude is dependent on the oscillated frequency. In this paper, a new current mode sinusoidal oscillator with a new technique to control amplitude is presented. The proposed circuit is based on translinear current controlled conveyors, consisting of 3 positive second generation current controlled
0-7803-8251-X/04/$17.00 ©2004 IEEE
conveyors (CCCII+) and 2 grounded capacitors (benificial to an IC implementation [8]). The outstanding features of the proposed circuit are that; it can simultaneously provide quadrature sinusoidal signals in both voltage and current modes, which is very profitable in communication systems; the oscillation frequency can be controlled by the input bias currents; and the quadrature amplitudes can be linearly controlled by another input bias current whereas no effect is on the condition of oscillation [9-10]. With the proposed principle, it can generate AM and FM signal without an additional device requirement. The PSPICE simulation results are also shown, which are in correspondence with the theoretical analysis. 2. CIRCUIT PRINCIPLE 2.1. The second-generation current controlled conveyor Since the proposed circuit is based on CCCIIs, a briefly review of CCCII is given in this section. Basically, the CCCII is composed of translinear elements, mixed loops and complementary current mirrors [7]. Generally, its properties are similar to the conventional CCII, except that the CCCII has a finite input resistance RX at the X terminal. This intrinsic resistance can be controlled by the bias current I B as shown in the following equation VT
=
RX
(1)
2I B
IB IX
X Z
IZ
Y (a)
X
IB
IX
IZ Z
RX
1
IX
Y (b) Fig. 1 Second generation current controlled conveyor (a) Symbol (b) Equivalent circuit
IV - 549
ISCAS 2004
➡
➡ is the thermal voltage. The symbol and the
From eqn. (10), the response of vo 3 (t ) can be decomposed into
equivalent circuit of the CCCII+ are illustrated in Fig. 1(a) and (b), respectively.
2 parts which are natural response: vo 3n (t ) and forced response:
2.2. The proposed circuit and operation Fig. 2 demonstrates the circuit scheme of quadrature oscillator. The depicted bias currents I B 1 , I B 2 and I B 3 are respectively
vo 3 (t ) = vo 3n (t ) + vo 3 f (t )
Where VT
the input bias currents of CCCII 1, CCCII 2 and CCCII 3. From the circuit and the CCCII properties in section 2.1, The following results are obtained Io1 (t ) = Io 3 (t ) Io2 (t ) = Io 3 (t ) =
(3)
RX 1 vo 2 (t ) RX 2 d v (t ) dt o2
I B1
v (t )¯ ¡¢ o 3 °±
C1
1
(7)
Y
X
I o 2 (t ) C2
2
Z
(14)
vo 3 (t ) = A cos Xot + B sin Xot
(16)
A and B coefficients can be determined by using the initial condition: vo 3 (0) = VSAT . From eqn. (16), we receive
X vo 3 (t )
CCCII +
2I 1 = B RXC VTC
The forced response can be subsequently calculated from eqn. (10) as follows vo 3 f (t ) = 0 (15)
I o 3 (t )
CCCII +
A = VSAT and B = 0
X Z
(13)
Consequently, the total output response; vo 3 (t ) can be shown as
I B2
Z
CCCII +
vo 3n (t ) = A cos Xot + B sin Xot
Xo =
(6)
vo 2 (t )
Y I o1 (t )
(12)
Then,
(5)
d2 vo 3 (t ) = RX 1RX 3C 1C 2 2 vo 2 (t ) dt vo 1 (t )
d2 ¯ ¡v (t )° + vo 3 (t ) = 0 dt 2 ¢ o 3 ±
RX2 C 2
While Xo is oscillation frequency equal to 1/ RXC or
vo1 (t ) = RX 1C 2 RX 3
is a
(4)
and
vo2 (t ) =
Whereas the initial oscillation: vo 3 (0) = VSAT , VSAT
(11)
saturation voltage of the CCCII dependent on power supply voltage. The natural response can be obtained by employing eqn. (12)
(2)
vo1 (t )
RX 2
vo 3 f (t ) as follows [9-10]
(17)
Thus, the completed output responses are
3 Y
IB3
vo 3 (t ) = (VSAT ) cos Xot
(18)
IB 3
(19)
vo2 (t ) =
IB
(VSAT ) cos Xot
Fig. 2 Block diagram of the proposed circuit
Similarly, substituting eqn. (19) into eqn. (5), then it yields By substituting eqn. (6) into (7), it can be shown that d2 vo 3 (t ) = RX 1RX 2C 1C 2 2 ¡¢vo 3 (t )i ¯°± dt
(8)
Thus, RX 1RX 2C 1C 2
d2 ¯ ¡v (t )° + vo 3 (t ) = 0 dt 2 ¢ o 3 ±
vo1 (t ) =
(VSAT ) cos (Xot + 90°)
(20)
Io2 (t ) can be obtained by using eqn. (2)-(4) as follows
as C and I B , respectively). The eqn. (9) becomes d2 ¯ ¡v (t )° + vo 3 (t ) = 0 dt 2 ¢ o 3 ±
IB
The quadrature sinusoidal signals in current mode: Io1 (t ) and (9)
If C 1 and C 2 , I B 1 and I B 2 are forced to be identical (designed
RX2 C 2
IB3
(10)
IV - 550
Io1 (t ) =
Io2 (t ) =
2I B 3 VT 2I B 3 VT
(VSAT ) sin (Xot 90°)
(21)
(VSAT ) cos (Xot + 90°)
(22)
➡
➡ 107
I B 3 is constant and I B is used as a modulating signal, the
106
Similarly, if I B is constant whereas I B 3 is a modulating signal, the quadrature AM signals can be accomplished at either Io1 (t ) and Io2 (t ) or vo1 (t ) and vo2 (t ) . 3. SIMULATED AND EXPERIENTAL RESULTS To prove the performances of the proposed circuit, the PSPICE simulation program was used for the examination. The PNP and NPN transistors employed in the proposed circuit were simulated by respectively using the parameters of the PR200N and NR200N bipolar transistors of ALA400 transistor array from AT&T [11]. Fig. 3 depicts schematic description of the CCCII+ used in the simulations. The power supply voltage was set at ±5V and the capacitors C 1 and C 2 are 10nF. Firstly, the
simulation results are in accordance with the theoretical analysis shown in eqn. (14). VCC Q8 Q10 Q7 Q9
104 103
101 10-7
10-6
10-5
10-4
10-3
Input Bias Currents (A) Fig. 5 Oscillation frequencies against bias currents of CCCIIs in the proposed circuit 8x10-3 Amplitude of current: Io2(t), (A)
depicts the plots of the simulated and theoretical oscillation frequency versus the bias currents: I B 1 , I B 2 . It is seen that the
105
102
input bias currents: I B 1 , I B 2 and I B 3 are respectively set to 150µA , 150µA and 5µA where are shown in Fig. 4. Fig. 5
C=10nF Theory C = 1nF Theory
800x10-3 Theory of Io2 amplitude
7x10-3 6x10
Simulated Amplitude of Io2(t) Theory af Vo2 amplitude Simulated Amplitude of Vo2(t)
-3
5x10-3
600x10-3
4x10-3
400x10-3
3x10-3 2x10-3 1x10
200x10-3
-3
0
Q1 Y
2
4
6
8
10
12
14
16
18
20
Input bias current: IB3 (PA)
Z
X Q3
IB
0 0
Q2
Amplitude of voltage: Vo2(t), (V)
quadrature FM signals can be obtained at Io1 (t ) and Io2 (t ) .
Frequency (Hz)
From eqns. (19) through (22), they should be noted that the amplitude of sinusoidal signal in current-mode and voltagemode can be proportionally controlled by I B 3 . In addition, if
Q4
Fig. 6 Amplitudes of v02 (t ) and I 02 (t ) against bias current: I B 3
Q13
Q13
Q5
Q6 VEE
Fig. 3 Schematic description of the CCCII+
Total Harmonic Distortion(THD)
.08
Q12
.06
.04
.02
0.00
0.0
.2
.4
.6
.8
1.0
Frequency(MHz)
Fig. 7 The simulated result of THD with 10nF capacitors
The simulated results of amplitude control by adjusting I B 3 are Fig. 4 The simulation results of waveforms in the circuit during steady stage Top traces: , Io1 (t ), Io 2 (t ) Bottom traces: , vo1 (t ),
vo 2 (t )
shown in Fig.6. In this case, the input bias currents: I B 1 and I B 2 were set to be 200µA . As predicted from eqns. (19)-(22), the simulated results are in correspondence with the expectation. Fig. 7 insists that the simulated result of THD (Total Harmonic Distortion) is less than 7% with 10nF capacitors. The simulated
IV - 551
➡
➠ results of the proposed circuit serving as an FM and an AM signal generator are also illustrated in Fig. 8 and Fig. 9, respectively. They exhibit a good agreement between the theoretical analysis as shown in eqns. (19)-(22) and the simulation. In addition, to insist that the proposed circuit can operate practically, an experimental result of the proposed circuit employing CA3096s transistor arrays as discrete elements is also depicted in Fig. 10.
4. CONCLUSIONS The quadrature sinusoidal oscillator with amplitude control has been presented. The quadrature signals can be utilized in either current mode or voltage mode. The oscillated frequency can be tuned by adjusting I B 1 and I B 2 over a wide frequency range. Furthermore, the amplitude control can also be achieved by tuning another input bias current: I B 3 . The proposed circuit can generate either quadrature AM or FM signals without an additional element requirement. All of simulation experimental results confirm the theoretical analysis. Since the proposed circuit consists of merely three CCCIIs and two grounded capacitors, it is suitable for monolithic implementation in IC technology. The proposed circuit is expected to be useful for applications in communication, instrumentation and measurement systems, especially at a wide range of frequencies.
5. REFERENCES [1]
Fig. 8 The result of operation as FM signal generator where I B 3 = 5µA;C = 5nF
Fig. 9 The result of operation as AM signal generator where I B 1 = I B 2 = 50µA;C = 5nF
Liu S. I., and Tsay J. H.. “Single-resistance-controlled sinusoidal oscillator using current-feeback amplifiers”. Int. J. Electronics, 80:661-664, 1996. [2] Cicekoglu O., Cam U., and Kuntman H. “Single-resistancecontrolled sinusoidal oscillators employing single FTFN and grounded capacitors”, Proceedings of the 44th IEEE 2001 Midwest Symp. on Circuits and Systems, Dayton, OH, pages 874 –877. August 2001. [3] Filanovsky I. M., Qin S. S., and Kothapalli G. “Sinusoidal oscillator with voltage controlled frequency and amplitude”. Int. J. Electronics, 68:95-112, 1990. [4] Senani R., and Bhaskar D. R. “New active-R sinusoidal VCOs with linear tuning laws”. Int. J. Electronics, 80:57-61, 1996. [5] Pookaiyaudom S., and Saivichit K. “RC phase-shifter variable sinusoidal oscillators using initial conditions-restoration amplitude control”. IEEE Trans. on Instru & Meas., 39:10381044, 1990. [6] Kiranon W., Kesorn J., Sangpisit W., and Kamprasert N. “Electronically tunable multifunctional translinear-C filter and oscillator”. Electronics letters, 33:573-574, 1997. [7] Kiranon W., Kesorn J., and Wardkein P. “Current controlled oscillator based on translinear conveyors”. Electronics letters, 32:1330-1331, 1996. [8] Sun Y., and Fidler J.K. “Synthesis and performance analysis of universal minimum component integrator-based IFLF OTAgrounded capacitor filter”. IEE Proc. Circuit Devices Syst., 143:107-114, 1996. [9] Wardkein P., Wongsuwan C., and Maneechugate T. “The simple sinusoidal oscillator amplitude control by force response”. (in Thai), The 23rd EECON conference, Chaingmai, THAILAND, November 2000, pages 645-648. [10] Tammawut B., Wardkein P., and Maneechugate T. “The simple amplitude control by force response for current mode oscillator”. (in Thai), The 24th EECON conference, Bangkok, THAILAND, November 2001, pages 1018-1023. [11] Frey D. R. “Log-domain filtering: An approach to current-mode filtering”, IEE Proc. G., Circuits Devices Syst., 140:406-416, Dec. 1993.
! !
1) C h 1: 2) C h 2:
100 mVolt 5 us 100 mVolt 5 us
Fig. 10 Experimented result of quadrature output signals in voltage mode where I B 1 = I B 2 = 4.4mA, I B 3 = 0.88µA;C = 0.1µF
IV - 552
