Ultrabroadband Linear Power Amplifier Using a ... - Semantic Scholar
264
IEEE TRANSACTIONS ON CIRCUITS AND SYSTEMS—II: EXPRESS BRIEFS, VOL. 58, NO. 5, MAY 2011
Ultrabroadband Linear Power Amplifier Using a Frequency-Selective Analog Predistorter Mincheol Seo, Kyungwon Kim, Minsu Kim, Hyungchul Kim, Jeongbae Jeon, Myung-Kyu Park, Hyojoon Lim, and Youngoo Yang, Member, IEEE
Abstract—This brief presents an ultrabroadband (2- to 600-MHz band) linear power amplifier using a compact frequency-selective analog predistorter, which includes a capacitor having an optimized capacitance and a biased Schottky diode connected in series. It is shunted at the input of the main amplifier and has a frequency-selective third-order intermodulation generation capability using its optimized capacitance. A two-stage push–pull power amplifier, which has an ultrabroadband operation range from 2 to 600 MHz, was implemented and linearized using the proposed analog predistorter. It exhibited high 1-dB compression point (P1 dB) and power-added efficiency characteristics of over 43 dBm and 32% at each P1 dB for the entire operating band, respectively. At an average output power level of 36 dBm for the two-tone signal input, which has a tone spacing of 1 MHz, its third-order intermodulation distortion over the entire band is no higher than −39.1 dBc after linearization, as compared with −34.2 dBc before linearization. Index Terms—Analog predistorter, broadband power amplifier, frequency-selective predistorter, linearization, push–pull.
I. I NTRODUCTION
T
HERE are many kinds of broadband wireless communication systems or applications, such as military wireless communication systems, cable or satellite television services, and various medical and general-purpose applications. To achieve a higher data rate, some broadband systems are even required to employ a modulated signal with a nonconstant but time-varying envelope. To transmit these nonconstant envelope signals without considerable distortion, wireless communication systems require linear transmitting systems, in which the power amplifiers are the most critical parts [1]–[7]. For broadband amplifiers, which utilize the constant envelope signals generally adopted by the frequency- or phase-modulation scheme, unlike their harmonics, their intermodulation distortion characteristics have not been seriously considered. However, it is vital that they be sufficiently linear to utilize the modulated signals with a noncon-
Manuscript received May 28, 2010; revised August 19, 2010 and December 13, 2010; accepted March 6, 2011. Date of publication May 19, 2011; date of current version June 8, 2011. This work was supported by the National Research Foundation of Korea funded by the Korean Government under Grant 2009-0067097. This paper was recommended by Associate Editor E. Kerherve. M. Seo, K. Kim, M. Kim, H. Kim, J. Jeon, and Y. Yang are with the Microwave Circuits and Systems Laboratory, School of Information and Communication Engineering, Sungkyunkwan University, Suwon 440-746, South Korea (e-mail: [email protected]). M.-K. Park is with Peopleworks Inc., Seoul 152-766, South Korea. H. Lim is with LIG-Nex1 Co. Ltd., Yongin 446-912, South Korea. Digital Object Identifier 10.1109/TCSII.2011.2149170
stant envelope, such as 16-quadrature-amplitude modulation (16-QAM), 64-QAM, or quadrature phase-shift keying. Their harmonics can be mitigated by using a push–pull structure or additional cosite filter banks at the output of the amplifier, but not their intermodulation terms. To reduce nonlinearity, particularly the third-order intermodulation distortion (IMD3), various linearization techniques have been reported. Digital predistortion methods are the most powerful and popular ones [7]. They require a very high speed digital signal processing unit, as well as high-speed highresolution digital-to-analog and analog-to-digital converters, which are expensive and consume a significant amount of power. As the data rate or signal bandwidth increases, the speed of the signal processing and data conversion units should be increased. There are also many simple and inexpensive analog predistortion methods [1]–[6]. However, their performances are generally not comparable with those of digital predistortion methods. Furthermore, most of them have only been used in very narrow-band applications. In this brief, we propose a very simple analog predistorter for ultrabroadband power amplifiers. It is basically a shunt diode predistorter, which has frequency-selective characteristics that are obtained by using a series capacitor. Broadband power amplifiers exhibit nonlinear behavior in the sense that they are linear in the very low frequency band and become increasingly nonlinear as their operating frequency increases. Therefore, we can tune the predistorter circuit, particularly the value of the series capacitor, to operate in the frequency region in which the amplifier is highly nonlinear. Hence, the proposed predistorter generates almost no intermodulation terms in the very low frequency band while generating comparable antiphased intermodulation signals in the high-frequency band with a similar level to that of the main amplifier. A two-stage push–pull broadband amplifier, including the proposed compact frequency-selective analog predistorter, is designed and implemented. The output performances of the main amplifier, with and without the predistorter, are compared for the purpose of verification.
II. U LTRABROADBAND P OWER A MPLIFIER D ESIGN A. Main Amplifier Broadband amplifiers generally utilize a feedback method to flatten their power gain response over a wide operating frequency band [8]–[12]. Fig. 1 shows a schematic of the designed two-stage broadband amplifier, which has a feedback
1549-7747/$26.00 © 2011 IEEE
SEO et al.: ULTRABROADBAND POWER AMPLIFIER USING A FREQUENCY-SELECTIVE ANALOG PREDISTORTER
265
Fig. 3. Schematics of the shunt diode predistorters. (a) Conventional. (b) Proposed. Fig. 1. Schematic of the two-stage broadband amplifier using the feedback method.
Fig. 4. Simulated responses with various capacitance values of Copt of the proposed frequency-selective shunt diode predistorter. (a) IMD3. (b) S11. Fig. 2. Simulated characteristics of the two-stage broadband amplifier with and without feedback. (a) Gain. (b) IMD3.
network for each stage. The first-stage consists of a 10-W peak envelope power (PEP) Freescale MRF282 lateral doublediffused MOSFET (LDMOSFET), a series feedback network (inductor–resistor–capacitor) from the drain to the gate, and a broadband radio-frequency (RF) choke inductor to feed Vdd1. The second-stage consists of a Cree 40025 25-W PEP GaN high-electron mobility transistor (HEMT), a feedback network, and an RF choke inductor. The load and source impedance values of the amplifier are provided by the input and output baluns, which are used for push–pull operation [13]–[15]. The feedback network is used to flatten the power gain response over a wide frequency range, as shown in Fig. 2(a). The amplifier’s gain characteristics without feedback are very high,
particularly in the low-frequency band, and fall very sharply at a frequency of approximately 100 MHz. To make the gain flat, the feedback network, which is configured using lumped passive components connected in series, should be optimized to provide a large negative feedback at low frequency, resulting in a significant decrease of the gain. Otherwise, as the frequency increases, its feedback level would gradually decrease. Because the feedback improves its linearity, as well as the flatness of its gain, the IMD characteristics of the twostage broadband amplifier would be expected to worsen as the frequency increases, as shown in Fig. 2(b), which shows its simulated IMD3 characteristics with and without feedback at a two-tone average output power level of 33 dBm. The tone spacing for the two-tone excitation is 1 MHz. It is seen that the feedback clearly improves the IMD3 characteristics,
266
IEEE TRANSACTIONS ON CIRCUITS AND SYSTEMS—II: EXPRESS BRIEFS, VOL. 58, NO. 5, MAY 2011
Fig. 5. Simulated amplitude responses and phase differences for the upper and lower IMD3 terms of the main amplifier and predistorter. (a) Amplitude response. (b) Phase difference.
particularly in the lower frequency band below 100 MHz. Therefore, it is necessary for the significantly high IMD3 level in the high-frequency region to be improved. B. Frequency-Selective Analog Predistorter A simple shunt diode predistorter, as shown in Fig. 3(a), was introduced based on the one described in [6]. A properly biased shunt diode can generate IMD3 terms that are out of phase with and have similar amplitudes to those of the main amplifier. At its input and output, the predistorter has large dc blocking capacitors. Although its nonlinear behavior extends constant over a very wide frequency range, it can be used in narrow-band applications because the main amplifier’s nonlinear behavior as a function of frequency is nonconstant, as explained in the previous section. Fig. 3(b) shows the proposed frequency-selective shunt diode predistorter, which was simplified even further. It has a capacitor and a properly biased diode connected in series. Capacitor Copt has a finite capacitance, which endows the predistorter with frequency-selection properties. If we assume that the diode can be simply modeled as a nonlinear resistance Rdiode , the voltage across the diode becomes as follows: jωRdiode Copt VPD (1) Vdiode = 1 + jωRdiode Copt
Fig. 6.
Schematic of the linearized push–pull broadband power amplifier.
Fig. 7.
Photograph of the implemented broadband linear power amplifier.
where Vdiode and VPD are the voltages across the diode and the entire predistorter, respectively. In the case where Copt is close to infinity, Vdiode VPD , which means that there is no frequency-selection property in the predistorter. Otherwise, if Copt is a finite value, Vdiode has high-pass characteristics. The amplitude response of Vdiode over VPD increases by +20 dB/dec below the 3-dB frequency, which is 1/(Copt Rdiode ). As the voltage across the diode, i.e., Vdiode , increases according to the voltage applied across the entire predistorter, i.e., VPD , the IMD3 generated by the diode increases. Fig. 4(a) represents the generated IMD3 level as a function of frequency as the value of the shunt capacitor varies from 1.5 pF to infinity. Using the input impedance, we can derive the input reflection coefficient, which is the same as the value of S11 of the shunt predistorter, i.e., Zin = Rdiode + S11 =
1 jωCopt
1 + jω(Rdiode − Z0 )Copt Zin − Z0 = . Zin + Z0 1 + jω(Rdiode + Z0 )Copt
(2) (3)
Because |Rdiode − Z0 | < |Rdiode + Z0 |, S11 has low-pass characteristics whose 3-dB frequency of 1/(Copt (Rdiode + Z0 )) is much lower than that of Vdiode /VPD . Fig. 4(b) depicts the S11 responses for various values of Copt . Fig. 5(a) shows the simulated amplitude responses of the IMD3s that are generated by the main amplifier (solid lines) and the proposed predistorter (dotted lines). Using a Copt value of
SEO et al.: ULTRABROADBAND POWER AMPLIFIER USING A FREQUENCY-SELECTIVE ANALOG PREDISTORTER
267
Fig. 9. Measured gain, PAE, and P1 dB performances for the whole operating frequency band at an output power of 36 dBm. TABLE I S UMMARY OF THE M EASURED R ESULTS
III. I MPLEMENTATION AND E XPERIMENTAL R ESULTS
Fig. 8. Measured IMD3 characteristics with and without predistorter. (a) Below 100 MHz (power sweep). (b) Above 100 MHz (power sweep). (c) At an output power of 36 dBm (frequency sweep).
33 pF, the IMD3 amplitude of the predistorter tracks the IMD3 of the main amplifier sufficiently well above 100 MHz. Below 100 MHz, the IMD3 level of the predistorter quickly vanishes, so that it does not harm the original IMD3 characteristics of the main amplifier, which are already satisfactory. Fig. 5(b) shows the phase differences for the upper and lower IMD3 terms generated by the main amplifier and the predistorter. Since they are located near 180◦ in the frequency region above 100 MHz, good cancellation is to be expected.
Based on the design of the two-stage feedback main amplifier and the frequency-selective diode predistorter, the linearized broadband push–pull power amplifier, whose schematic is shown in Fig. 6, is assembled. The detailed schematic of the main amplifier is shown in Fig. 1. Skyworks SMS1546005 Schottky diodes are used for the analog predistorter. For the push–pull configuration, Guanella 1 : 1 transmission line transformers are adopted for both the input and the output [13]. The 1 : 1 transformer provides the load and the source of the individual two-stage amplifier with an impedance value of 25 Ω. It is implemented on a printed circuit board that has a size of 107 × 69 mm2 . A photograph of the implemented broadband power amplifier is shown in Fig. 7. It also includes the bias circuits that generate the negative gate voltage for the GaN HEMT devices. A drain voltage of 26 V with a quiescent current of 2.4 A is applied. Fig. 8 shows the measured IMD3 performances of the push–pull broadband amplifier with (lines) and without predistorter (lines and symbols). Fig. 8(a) and (b) shows the IMD3 performances below and above 100 MHz, respectively. In the low-frequency region, the IMD3 characteristics of the main amplifier with the predistorter (expressed as PD in the figure) are sufficiently good and are virtually unchanged compared with the original ones without the predistorter. On the contrary, the IMD3 characteristics in the high-frequency region are significantly improved. At an output power of 36 dBm, which corresponds to a back-off of about 7 dB from the 1-dB compression point (P1 dB), the worst IMD3 values are −39.1 and −34.dBc with and without the predistorter, respectively, as shown more clearly in Fig. 8(c).
268
IEEE TRANSACTIONS ON CIRCUITS AND SYSTEMS—II: EXPRESS BRIEFS, VOL. 58, NO. 5, MAY 2011
TABLE II P ERFORMANCE C OMPARISON W ITH P REVIOUS W ORKS
Fig. 9 presents the measured gain, power-added efficiency (PAE), and P1 dB performances of the linearized broadband power amplifier. The measured gain is as flat as ±1.3 dB over the band from 2 to 600 MHz. The power gain with the predistorter is nearly unchanged compared with that of the original circuit without the predistorter. The P1 dB and PAE values at each P1 dB over the entire band are no less than 43 dBm and 32%, respectively. A peak PAE value of 46.1% is observed at 100 MHz. Table I summarizes the measured results. Table II compares the results obtained from this work with those of other previously published works. Compared with [8] and [9], better IMD3 performance was achieved at the 7 dB back-off point with much higher PAE performance. The implemented broadband PA with the analog PD also shows comparable PAE performance with [10] and [11], whose IMD3 performances are not available. IV. C ONCLUSION The IMD3 characteristics of ultrabroadband amplifiers are generally a sharply increasing function of frequency because they utilize the negative feedback method to ensure the flatness of the gain. To improve their IMD3 characteristics, we have proposed a compact diode predistorter having frequency-selection properties, so that it can maximally cancel the IMD3 of the main amplifier in the high-frequency region while not affecting its good IMD3 in the low-frequency region. To validate the proposed predistorter, two two-stage power amplifiers were designed and implemented. They are balanced for two-way push–pull operation. Each two-stage amplifier was linearized using the proposed frequency-selective predistorter in the overall push–pull amplifier. The implemented ultrabroadband amplifier exhibited similar or better performance in terms of its operational bandwidth, gain, output power level, and efficiency, as compared with those of previous works. It also delivered excellent IMD3 performance after linearization. A 4.9-dB improvement in IMD3 was achieved at an output power of 36 dBm, which corresponds to a back-off of about 7 dB from the P1 dB. To the best of our knowledge, this is the first report of an ultrabroadband power amplifier having a multidecade
bandwidth that achieves successful linearization, particularly in terms of IMD3. The proposed predistorter can also be applied to broadband amplifiers based on GaAs field-effect transistors or silicon LDMOSFETs. R EFERENCES [1] S. Jung, H. Park, M. Kim, G. Ahn, J. Van, H. Hwangbo, C. Park, S. Park, and Y. Yang, “A new envelope predistorter with envelope delay taps for memory effect compensation,” IEEE Trans. Microw. Theory Tech., vol. 55, no. 1, pp. 52–59, Jan. 2007. [2] K. Lim, G. Ahn, S. Jung, H. Park, M. Kim, J. Van, H. Cho, J. Jeong, C. Park, and Y. Yang, “A 60 watt multi-carrier WCDMA power amplifier using an RF predistorter,” IEEE Trans. Circuits Syst. II, Exp. Briefs, vol. 56, no. 4, pp. 265–269, Apr. 2009. [3] J. Yi, Y. Yang, M. G. Park, W. W. Kang, and B. Kim, “Analog predistortion linearizer for high-power RF amplifiers,” IEEE Trans. Microw. Theory Tech., vol. 48, no. 12, pp. 2709–2713, Dec. 2000. [4] T. Nojima and T. Konno, “Cuber predistortion linearizer for relay equipment in 800 MHz band land mobile telephone system,” IEEE Trans. Veh. Technol., vol. VT-34, no. 4, pp. 169–177, Nov. 1985. [5] J. Cha, J. Yi, J. Kim, and B. Kim, “Optimum design of a predistortion RF power amplifier for multicarrier WCDMA applications,” IEEE Trans. Microw. Theory Tech., vol. 52, no. 2, pp. 655–663, Feb. 2004. [6] K. Yamauchi, K. Mori, M. Nakayama, Y. Mitsui, and T. Takagi, “A microwave miniaturized linearizer using a parallel diode,” in Proc. IEEE MTT-S Int. Microw. Symp. Dig., Jun. 1997, pp. 1199–1202. [7] Y. Y. Woo, J. Kim, J. Yi, S. Hong, I. Kim, J. Moon, and B. Kim, “Adaptive digital feedback predistortion technique for linearizing power amplifiers,” IEEE Trans. Microw. Theory Tech., vol. 55, no. 5, pp. 932– 940, May 2007. [8] A. K. Ezzeddine and H. C. Huang, “10 W ultra-broadband power amplifier,” in Proc. IEEE MTT-S Int. Microw. Symp. Dig., Jun. 2008, pp. 643–646. [9] A. K. Ezzeddine and H. C. Huang, “Ultra-broadband GaAs HIFET MMIC PA,” in Proc. IEEE MTT-S Int. Microw. Symp. Dig., Jun. 2006, pp. 1320–1323. [10] S. You, K. Lim, J. Cho, M. Seo, K. Kim, J. Sim, M. Park, and Y. Yang, “A 5W ultra-broadband power amplifier using silicon LDMOSFETs,” in Proc. Asia-Pacific Microw. Conf. Dig., Dec. 2009, pp. 1116–1119. [11] N. Sahan, M. E. Inal, S. Demir, and C. Toker, “High-power 20–100-MHz linear and efficient power-amplifier design,” IEEE Trans. Microw. Theory Tech., vol. 56, no. 9, pp. 2032–2039, Sep. 2008. [12] W. H. Lambert, “Second-order distortion in CATV push–pull amplifiers,” Proc. IEEE, vol. 58, no. 7, pp. 1057–1062, Jul. 1970. [13] G. Guanella, “Nouveau transformateur d¡¯adaptation pour haute frequence,” Rev. Brown Boveri, vol. 31, pp. 327–329, Sep. 1944. [14] P. Gomez-Jimenez, P. Otero, and E. Marquez-Segura, “Analysis and design procedure of transmission-line transformers,” IEEE Trans. Microw. Theory Tech., vol. 56, no. 1, pp. 163–171, Jan. 2008. [15] C. L. Ruthroff, “Some broadband transformers,” Proc. IRE, vol. 47, no. 8, pp. 1337–1342, Aug. 1959.
IEEE TRANSACTIONS ON CIRCUITS AND SYSTEMS—II: EXPRESS BRIEFS, VOL. 58, NO. 5, MAY 2011
Ultrabroadband Linear Power Amplifier Using a Frequency-Selective Analog Predistorter Mincheol Seo, Kyungwon Kim, Minsu Kim, Hyungchul Kim, Jeongbae Jeon, Myung-Kyu Park, Hyojoon Lim, and Youngoo Yang, Member, IEEE
Abstract—This brief presents an ultrabroadband (2- to 600-MHz band) linear power amplifier using a compact frequency-selective analog predistorter, which includes a capacitor having an optimized capacitance and a biased Schottky diode connected in series. It is shunted at the input of the main amplifier and has a frequency-selective third-order intermodulation generation capability using its optimized capacitance. A two-stage push–pull power amplifier, which has an ultrabroadband operation range from 2 to 600 MHz, was implemented and linearized using the proposed analog predistorter. It exhibited high 1-dB compression point (P1 dB) and power-added efficiency characteristics of over 43 dBm and 32% at each P1 dB for the entire operating band, respectively. At an average output power level of 36 dBm for the two-tone signal input, which has a tone spacing of 1 MHz, its third-order intermodulation distortion over the entire band is no higher than −39.1 dBc after linearization, as compared with −34.2 dBc before linearization. Index Terms—Analog predistorter, broadband power amplifier, frequency-selective predistorter, linearization, push–pull.
I. I NTRODUCTION
T
HERE are many kinds of broadband wireless communication systems or applications, such as military wireless communication systems, cable or satellite television services, and various medical and general-purpose applications. To achieve a higher data rate, some broadband systems are even required to employ a modulated signal with a nonconstant but time-varying envelope. To transmit these nonconstant envelope signals without considerable distortion, wireless communication systems require linear transmitting systems, in which the power amplifiers are the most critical parts [1]–[7]. For broadband amplifiers, which utilize the constant envelope signals generally adopted by the frequency- or phase-modulation scheme, unlike their harmonics, their intermodulation distortion characteristics have not been seriously considered. However, it is vital that they be sufficiently linear to utilize the modulated signals with a noncon-
Manuscript received May 28, 2010; revised August 19, 2010 and December 13, 2010; accepted March 6, 2011. Date of publication May 19, 2011; date of current version June 8, 2011. This work was supported by the National Research Foundation of Korea funded by the Korean Government under Grant 2009-0067097. This paper was recommended by Associate Editor E. Kerherve. M. Seo, K. Kim, M. Kim, H. Kim, J. Jeon, and Y. Yang are with the Microwave Circuits and Systems Laboratory, School of Information and Communication Engineering, Sungkyunkwan University, Suwon 440-746, South Korea (e-mail: [email protected]). M.-K. Park is with Peopleworks Inc., Seoul 152-766, South Korea. H. Lim is with LIG-Nex1 Co. Ltd., Yongin 446-912, South Korea. Digital Object Identifier 10.1109/TCSII.2011.2149170
stant envelope, such as 16-quadrature-amplitude modulation (16-QAM), 64-QAM, or quadrature phase-shift keying. Their harmonics can be mitigated by using a push–pull structure or additional cosite filter banks at the output of the amplifier, but not their intermodulation terms. To reduce nonlinearity, particularly the third-order intermodulation distortion (IMD3), various linearization techniques have been reported. Digital predistortion methods are the most powerful and popular ones [7]. They require a very high speed digital signal processing unit, as well as high-speed highresolution digital-to-analog and analog-to-digital converters, which are expensive and consume a significant amount of power. As the data rate or signal bandwidth increases, the speed of the signal processing and data conversion units should be increased. There are also many simple and inexpensive analog predistortion methods [1]–[6]. However, their performances are generally not comparable with those of digital predistortion methods. Furthermore, most of them have only been used in very narrow-band applications. In this brief, we propose a very simple analog predistorter for ultrabroadband power amplifiers. It is basically a shunt diode predistorter, which has frequency-selective characteristics that are obtained by using a series capacitor. Broadband power amplifiers exhibit nonlinear behavior in the sense that they are linear in the very low frequency band and become increasingly nonlinear as their operating frequency increases. Therefore, we can tune the predistorter circuit, particularly the value of the series capacitor, to operate in the frequency region in which the amplifier is highly nonlinear. Hence, the proposed predistorter generates almost no intermodulation terms in the very low frequency band while generating comparable antiphased intermodulation signals in the high-frequency band with a similar level to that of the main amplifier. A two-stage push–pull broadband amplifier, including the proposed compact frequency-selective analog predistorter, is designed and implemented. The output performances of the main amplifier, with and without the predistorter, are compared for the purpose of verification.
II. U LTRABROADBAND P OWER A MPLIFIER D ESIGN A. Main Amplifier Broadband amplifiers generally utilize a feedback method to flatten their power gain response over a wide operating frequency band [8]–[12]. Fig. 1 shows a schematic of the designed two-stage broadband amplifier, which has a feedback
1549-7747/$26.00 © 2011 IEEE
SEO et al.: ULTRABROADBAND POWER AMPLIFIER USING A FREQUENCY-SELECTIVE ANALOG PREDISTORTER
265
Fig. 3. Schematics of the shunt diode predistorters. (a) Conventional. (b) Proposed. Fig. 1. Schematic of the two-stage broadband amplifier using the feedback method.
Fig. 4. Simulated responses with various capacitance values of Copt of the proposed frequency-selective shunt diode predistorter. (a) IMD3. (b) S11. Fig. 2. Simulated characteristics of the two-stage broadband amplifier with and without feedback. (a) Gain. (b) IMD3.
network for each stage. The first-stage consists of a 10-W peak envelope power (PEP) Freescale MRF282 lateral doublediffused MOSFET (LDMOSFET), a series feedback network (inductor–resistor–capacitor) from the drain to the gate, and a broadband radio-frequency (RF) choke inductor to feed Vdd1. The second-stage consists of a Cree 40025 25-W PEP GaN high-electron mobility transistor (HEMT), a feedback network, and an RF choke inductor. The load and source impedance values of the amplifier are provided by the input and output baluns, which are used for push–pull operation [13]–[15]. The feedback network is used to flatten the power gain response over a wide frequency range, as shown in Fig. 2(a). The amplifier’s gain characteristics without feedback are very high,
particularly in the low-frequency band, and fall very sharply at a frequency of approximately 100 MHz. To make the gain flat, the feedback network, which is configured using lumped passive components connected in series, should be optimized to provide a large negative feedback at low frequency, resulting in a significant decrease of the gain. Otherwise, as the frequency increases, its feedback level would gradually decrease. Because the feedback improves its linearity, as well as the flatness of its gain, the IMD characteristics of the twostage broadband amplifier would be expected to worsen as the frequency increases, as shown in Fig. 2(b), which shows its simulated IMD3 characteristics with and without feedback at a two-tone average output power level of 33 dBm. The tone spacing for the two-tone excitation is 1 MHz. It is seen that the feedback clearly improves the IMD3 characteristics,
266
IEEE TRANSACTIONS ON CIRCUITS AND SYSTEMS—II: EXPRESS BRIEFS, VOL. 58, NO. 5, MAY 2011
Fig. 5. Simulated amplitude responses and phase differences for the upper and lower IMD3 terms of the main amplifier and predistorter. (a) Amplitude response. (b) Phase difference.
particularly in the lower frequency band below 100 MHz. Therefore, it is necessary for the significantly high IMD3 level in the high-frequency region to be improved. B. Frequency-Selective Analog Predistorter A simple shunt diode predistorter, as shown in Fig. 3(a), was introduced based on the one described in [6]. A properly biased shunt diode can generate IMD3 terms that are out of phase with and have similar amplitudes to those of the main amplifier. At its input and output, the predistorter has large dc blocking capacitors. Although its nonlinear behavior extends constant over a very wide frequency range, it can be used in narrow-band applications because the main amplifier’s nonlinear behavior as a function of frequency is nonconstant, as explained in the previous section. Fig. 3(b) shows the proposed frequency-selective shunt diode predistorter, which was simplified even further. It has a capacitor and a properly biased diode connected in series. Capacitor Copt has a finite capacitance, which endows the predistorter with frequency-selection properties. If we assume that the diode can be simply modeled as a nonlinear resistance Rdiode , the voltage across the diode becomes as follows: jωRdiode Copt VPD (1) Vdiode = 1 + jωRdiode Copt
Fig. 6.
Schematic of the linearized push–pull broadband power amplifier.
Fig. 7.
Photograph of the implemented broadband linear power amplifier.
where Vdiode and VPD are the voltages across the diode and the entire predistorter, respectively. In the case where Copt is close to infinity, Vdiode VPD , which means that there is no frequency-selection property in the predistorter. Otherwise, if Copt is a finite value, Vdiode has high-pass characteristics. The amplitude response of Vdiode over VPD increases by +20 dB/dec below the 3-dB frequency, which is 1/(Copt Rdiode ). As the voltage across the diode, i.e., Vdiode , increases according to the voltage applied across the entire predistorter, i.e., VPD , the IMD3 generated by the diode increases. Fig. 4(a) represents the generated IMD3 level as a function of frequency as the value of the shunt capacitor varies from 1.5 pF to infinity. Using the input impedance, we can derive the input reflection coefficient, which is the same as the value of S11 of the shunt predistorter, i.e., Zin = Rdiode + S11 =
1 jωCopt
1 + jω(Rdiode − Z0 )Copt Zin − Z0 = . Zin + Z0 1 + jω(Rdiode + Z0 )Copt
(2) (3)
Because |Rdiode − Z0 | < |Rdiode + Z0 |, S11 has low-pass characteristics whose 3-dB frequency of 1/(Copt (Rdiode + Z0 )) is much lower than that of Vdiode /VPD . Fig. 4(b) depicts the S11 responses for various values of Copt . Fig. 5(a) shows the simulated amplitude responses of the IMD3s that are generated by the main amplifier (solid lines) and the proposed predistorter (dotted lines). Using a Copt value of
SEO et al.: ULTRABROADBAND POWER AMPLIFIER USING A FREQUENCY-SELECTIVE ANALOG PREDISTORTER
267
Fig. 9. Measured gain, PAE, and P1 dB performances for the whole operating frequency band at an output power of 36 dBm. TABLE I S UMMARY OF THE M EASURED R ESULTS
III. I MPLEMENTATION AND E XPERIMENTAL R ESULTS
Fig. 8. Measured IMD3 characteristics with and without predistorter. (a) Below 100 MHz (power sweep). (b) Above 100 MHz (power sweep). (c) At an output power of 36 dBm (frequency sweep).
33 pF, the IMD3 amplitude of the predistorter tracks the IMD3 of the main amplifier sufficiently well above 100 MHz. Below 100 MHz, the IMD3 level of the predistorter quickly vanishes, so that it does not harm the original IMD3 characteristics of the main amplifier, which are already satisfactory. Fig. 5(b) shows the phase differences for the upper and lower IMD3 terms generated by the main amplifier and the predistorter. Since they are located near 180◦ in the frequency region above 100 MHz, good cancellation is to be expected.
Based on the design of the two-stage feedback main amplifier and the frequency-selective diode predistorter, the linearized broadband push–pull power amplifier, whose schematic is shown in Fig. 6, is assembled. The detailed schematic of the main amplifier is shown in Fig. 1. Skyworks SMS1546005 Schottky diodes are used for the analog predistorter. For the push–pull configuration, Guanella 1 : 1 transmission line transformers are adopted for both the input and the output [13]. The 1 : 1 transformer provides the load and the source of the individual two-stage amplifier with an impedance value of 25 Ω. It is implemented on a printed circuit board that has a size of 107 × 69 mm2 . A photograph of the implemented broadband power amplifier is shown in Fig. 7. It also includes the bias circuits that generate the negative gate voltage for the GaN HEMT devices. A drain voltage of 26 V with a quiescent current of 2.4 A is applied. Fig. 8 shows the measured IMD3 performances of the push–pull broadband amplifier with (lines) and without predistorter (lines and symbols). Fig. 8(a) and (b) shows the IMD3 performances below and above 100 MHz, respectively. In the low-frequency region, the IMD3 characteristics of the main amplifier with the predistorter (expressed as PD in the figure) are sufficiently good and are virtually unchanged compared with the original ones without the predistorter. On the contrary, the IMD3 characteristics in the high-frequency region are significantly improved. At an output power of 36 dBm, which corresponds to a back-off of about 7 dB from the 1-dB compression point (P1 dB), the worst IMD3 values are −39.1 and −34.dBc with and without the predistorter, respectively, as shown more clearly in Fig. 8(c).
268
IEEE TRANSACTIONS ON CIRCUITS AND SYSTEMS—II: EXPRESS BRIEFS, VOL. 58, NO. 5, MAY 2011
TABLE II P ERFORMANCE C OMPARISON W ITH P REVIOUS W ORKS
Fig. 9 presents the measured gain, power-added efficiency (PAE), and P1 dB performances of the linearized broadband power amplifier. The measured gain is as flat as ±1.3 dB over the band from 2 to 600 MHz. The power gain with the predistorter is nearly unchanged compared with that of the original circuit without the predistorter. The P1 dB and PAE values at each P1 dB over the entire band are no less than 43 dBm and 32%, respectively. A peak PAE value of 46.1% is observed at 100 MHz. Table I summarizes the measured results. Table II compares the results obtained from this work with those of other previously published works. Compared with [8] and [9], better IMD3 performance was achieved at the 7 dB back-off point with much higher PAE performance. The implemented broadband PA with the analog PD also shows comparable PAE performance with [10] and [11], whose IMD3 performances are not available. IV. C ONCLUSION The IMD3 characteristics of ultrabroadband amplifiers are generally a sharply increasing function of frequency because they utilize the negative feedback method to ensure the flatness of the gain. To improve their IMD3 characteristics, we have proposed a compact diode predistorter having frequency-selection properties, so that it can maximally cancel the IMD3 of the main amplifier in the high-frequency region while not affecting its good IMD3 in the low-frequency region. To validate the proposed predistorter, two two-stage power amplifiers were designed and implemented. They are balanced for two-way push–pull operation. Each two-stage amplifier was linearized using the proposed frequency-selective predistorter in the overall push–pull amplifier. The implemented ultrabroadband amplifier exhibited similar or better performance in terms of its operational bandwidth, gain, output power level, and efficiency, as compared with those of previous works. It also delivered excellent IMD3 performance after linearization. A 4.9-dB improvement in IMD3 was achieved at an output power of 36 dBm, which corresponds to a back-off of about 7 dB from the P1 dB. To the best of our knowledge, this is the first report of an ultrabroadband power amplifier having a multidecade
bandwidth that achieves successful linearization, particularly in terms of IMD3. The proposed predistorter can also be applied to broadband amplifiers based on GaAs field-effect transistors or silicon LDMOSFETs. R EFERENCES [1] S. Jung, H. Park, M. Kim, G. Ahn, J. Van, H. Hwangbo, C. Park, S. Park, and Y. Yang, “A new envelope predistorter with envelope delay taps for memory effect compensation,” IEEE Trans. Microw. Theory Tech., vol. 55, no. 1, pp. 52–59, Jan. 2007. [2] K. Lim, G. Ahn, S. Jung, H. Park, M. Kim, J. Van, H. Cho, J. Jeong, C. Park, and Y. Yang, “A 60 watt multi-carrier WCDMA power amplifier using an RF predistorter,” IEEE Trans. Circuits Syst. II, Exp. Briefs, vol. 56, no. 4, pp. 265–269, Apr. 2009. [3] J. Yi, Y. Yang, M. G. Park, W. W. Kang, and B. Kim, “Analog predistortion linearizer for high-power RF amplifiers,” IEEE Trans. Microw. Theory Tech., vol. 48, no. 12, pp. 2709–2713, Dec. 2000. [4] T. Nojima and T. Konno, “Cuber predistortion linearizer for relay equipment in 800 MHz band land mobile telephone system,” IEEE Trans. Veh. Technol., vol. VT-34, no. 4, pp. 169–177, Nov. 1985. [5] J. Cha, J. Yi, J. Kim, and B. Kim, “Optimum design of a predistortion RF power amplifier for multicarrier WCDMA applications,” IEEE Trans. Microw. Theory Tech., vol. 52, no. 2, pp. 655–663, Feb. 2004. [6] K. Yamauchi, K. Mori, M. Nakayama, Y. Mitsui, and T. Takagi, “A microwave miniaturized linearizer using a parallel diode,” in Proc. IEEE MTT-S Int. Microw. Symp. Dig., Jun. 1997, pp. 1199–1202. [7] Y. Y. Woo, J. Kim, J. Yi, S. Hong, I. Kim, J. Moon, and B. Kim, “Adaptive digital feedback predistortion technique for linearizing power amplifiers,” IEEE Trans. Microw. Theory Tech., vol. 55, no. 5, pp. 932– 940, May 2007. [8] A. K. Ezzeddine and H. C. Huang, “10 W ultra-broadband power amplifier,” in Proc. IEEE MTT-S Int. Microw. Symp. Dig., Jun. 2008, pp. 643–646. [9] A. K. Ezzeddine and H. C. Huang, “Ultra-broadband GaAs HIFET MMIC PA,” in Proc. IEEE MTT-S Int. Microw. Symp. Dig., Jun. 2006, pp. 1320–1323. [10] S. You, K. Lim, J. Cho, M. Seo, K. Kim, J. Sim, M. Park, and Y. Yang, “A 5W ultra-broadband power amplifier using silicon LDMOSFETs,” in Proc. Asia-Pacific Microw. Conf. Dig., Dec. 2009, pp. 1116–1119. [11] N. Sahan, M. E. Inal, S. Demir, and C. Toker, “High-power 20–100-MHz linear and efficient power-amplifier design,” IEEE Trans. Microw. Theory Tech., vol. 56, no. 9, pp. 2032–2039, Sep. 2008. [12] W. H. Lambert, “Second-order distortion in CATV push–pull amplifiers,” Proc. IEEE, vol. 58, no. 7, pp. 1057–1062, Jul. 1970. [13] G. Guanella, “Nouveau transformateur d¡¯adaptation pour haute frequence,” Rev. Brown Boveri, vol. 31, pp. 327–329, Sep. 1944. [14] P. Gomez-Jimenez, P. Otero, and E. Marquez-Segura, “Analysis and design procedure of transmission-line transformers,” IEEE Trans. Microw. Theory Tech., vol. 56, no. 1, pp. 163–171, Jan. 2008. [15] C. L. Ruthroff, “Some broadband transformers,” Proc. IRE, vol. 47, no. 8, pp. 1337–1342, Aug. 1959.
