A 3.5GHz Low Power Programmable Transversal ... - Semantic Scholar
IEEE BCTM 4.4
A 3.5GHz Low Power Programmable Transversal Filter RFIC Implemented in 47GHz SiGe Technology Vasanth Kakani, Xuefeng Yu, Foster F. Dai, Richard C. Jaeger
Department of Electrical and Computer Engineering Auburn University, Auburn, AL 36849-5201, USA
II.
Abstract This paper presents the design of a low power 3.5GHz analog programmable filter RFIC. The RF filter is a 7-tap transversal equalizer with cascaded Cherry-Hooper amplifiers for delay stages and Gilbert variable gain amplifier as tap weights. The delay stage using active devices greatly reduces the die area comparing to passive delay lines. The SiGe programmable filter RFIC consumes 250mW under 3.3V supply and occupies total 2.16mm2 die size.
Index Terms SiGe, transversal filter, analog filter, Cherry-Hooper amplifier.
TRANSVERSAL FILTER DESIGN
The block diagram of the implemented transversal filter is shown in Fig. 1. The RF filter includes an analog tapped delay line with the feed forward taps forming an FIR filter. The transfer function of the integrated filter can be adaptively adjusted by changing its tap weights. Changing the tap weights affects only the locations of the zero's, while the poles of the programmable filter are fixed. Hence the filter is always stable.
Filter Input
I. INTRODUCTION
Programmable RF filters find numerous applications in communication systems. In fiber communications, modal, chromatic and polarization mode dispersions are the major sources of transmission impairments. Electronic transversal filters can be used to compensate fiber dispersions by constructing an inverse transfer function of the dispersive channel [1]. For multi-band wireless transceiver designs, programmable RF notch filters are needed to selectively reject the bands based on various wireless standards. RF notch filters are critical for removing unwanted signals such as images and interferers. In [2], a fractionally spaced transversal filter is designed using passive transmission lines as delay elements. Passive delay elements occupy large die area and provide accurate delays only for a narrow frequency band. In this paper, we present a 3.5GHz programmable filter RFIC designed in 45GHz SiGe technology. In stead of using passive delay lines, we propose to use Cherry-Hooper amplifiers as the delay stages, which provide wide bandwidth and occupy small area. We used Gilbert variable gain amplifier for continuous tuning of the tap weights. Measured S21 of the programmable RF filter demonstrates various filter characteristics such low-pass, high pass and notch filters up to 3.5GHz frequency.
1-4244-0459-2/06/$20.00 ©2006 IEEE.
Filter Output Fig. 1. Block diagram of the transversal filter. T denotes the delay amplifier and ® denotes a variable gain amplifier.
Passive delay networks are either lossy (RC delay lines) or bulky (LC delay lines). Moreover, passive delay networks are always narrow band. In stead of using passive delay stages, we chose the series shunt cascaded Cherry-Hooper amplifier to implement the filter delay stages. As shown in Fig. 2, Cherry-Hooper amplifier [3] is a cascade of two feedback amplifiers, where the series feedback stage is a transconductance amplifier and shunt feedback stage is a trans-impedance amplifier. The amplifier with series feedback is driven from a low resistance voltage input obtained from the output of the shunt feedback stage. Conversely, the amplifier with shunt feedback is driven from a high resistance current input obtained from the output of the serial feedback stage. This arrangement is advantageous since the impedance requirement will be automatically
IEEE BCTM 4.4
satisfied at the input and output of the amplifier while cascading several such delay stages. Emitter followers are used in between the delay stages for level shifting and creating stronger impedance mismatch between succeeding stages to improve the bandwidth. Referred to Fig.2, the transfer function of the CherryHooper amplifier can be approximated as
k)O+SCF RI +g, -RF) Vi, 9,,3(1+S CF RF)(+S CE RE +9ml RE)
kOUI =g('+S CE
(1)
where RE and CE are the degeneration resistance and capacitance, respectively; RF and CF are the shunt feedback resistance and capacitance, respectively. The degeneration and feedback capacitors CE and CF introduce zeros in the frequency response and thereby maximize the amplifier bandwidth. The first zero of the amplifier frequency response is created by the degeneration capacitor CE, and the second zero is generated by the shunt feedback capacitor CF. Fig. 3 gives the simulated magnitude response of the amplifier. As shown, the 3-dB cutoff frequency of the amplifier is about 9 GHz.
Fig. 3 Simulated magnitude response of the delay stage with 3-dB bandwidth at 9 GHz
Figure 5 shows the small-signal model of the gain control circuit in the Gilbert cell. The Gilbert cell is a current amplifier with the current gain transfer function given as
I(S) __t9(1+SCIr.)-g,(1+SC,,3r,)l {glni r3+ss
g.4 (2)
RE
dc
RE
Fig. 4 Tap weights implemented using Gilbert variable gain amplifier. Fig. 2 Cherry--Hooper amplifier used to implement delay stages.
As shown in Fig. 4, the transversal filter tap with programmable gain is implemented using Gilbert variable gain amplifier. Tap weights are continuously adjustable between 0 and 1 of the CML logic level (about +100mV differential). Thus the variable gain stage is infact a variable loss stage. Flipping the polarity of the gain control signal Vage provides a phase shift of 1800 for negative tap coefficients.
Figure 6 shows the simulated magnitude response of the Gilbert variable gain amplifier. The 3-dB cutoff frequency is achieved at 14.5 GHz.
Finally the summation required in the transversal filter is performed in the current mode. The output current signals of all the taps are tied together to an external pull up resistor via a current buffer, which is a common-base amplifier.
IEEE BCTM 4.4
tati'at
constant current sources for the amplifiers. The filter RFIC also includes an input buffer, output buffers and CMOS to CML buffers. Figure 7 shows the die photo of the transversal filter RFIC.
The frequency response of the integrated filter was measured using a vector network analyzer. The measured filter transfer functions under different tap weights are shown in Fig. 8 to Fig. 11. lF~
-b3 l-
lin
-10
Fig. 5 Small-signal model of the gain control circuit.
--------------
-15 -26
0.0 I-, (NJ
U)
-1 .0
(D-
-30
-40
---
--
,15
-4,0 -50
0.00
5.00G
10AG
freq ( Hz )
Fig.6 Simulated magnitude response of the Gilbert variable gain amplifier. The 3-dB frequency is 14.5 GHz.
II
1
15
2
*-
2.5
45
4
3
FREQUENCY
x 100
Fig. 8 Measured filter transfer function with double notches at 2.3GHz and 3.3 GHz. The tap coefficients are set as -40, 75, -40, 75, -40, 75, 90 [mV]. The magnitude of the notch is -55dB, which provides -37dB notch rejection compared to the pass band magnitude.
IV. MEASURED RESULTS 10
--
-15
30~--
1
-- - --
f
The 3.5 GHz transversal filter is implemented in a 45GHz SiGe technology with a total 2.16mm2 die area including pads. The chip consumes 250mW power under a 3.3V supply voltage. The filter RFIC includes a bandgap reference to provide temperature independent
--E --
_
-n- -- -- - -- r- -- -- - -- -n-- ----- ir-- -- - -- -,--- -- - -- -T- -- -- - -- -1--- -- - -- -T-- -- -- ---
FREQUENCY Fig. 7. Die photo of the integrated transversal filter RFIC.
-----
x
loo
Fig. 9 Measured filter transfer function with notch at 2GHz. The filter coefficients are set as -85, 30, -20, 0, 30, 0, 0 [mV]. The magnitude of the notch is -43db, which provides -30db notch rejection compared to the pass band magnitude.
IEEE BCTM 4.4
notch rejection compared to the pass band magnitude. Thus, the integrated programmable RF filter is able to tune zeros to any frequency up to 3.5 GHz. V. CONCLUSIONS
We have implemented a low power 3.5GHz analog transversal filter in a 47GHz SiGe technology. The RF filter utilizes cascaded Cherry-Hooper amplifiers for delay stages and Gilbert variable gain amplifier as for continuous gain tuning. The delay stage using active devices greatly reduces the die area comparing to passive delay lines. Measured results show that by adjusting the tap coefficients, the integrated programmable filter IC is capable to adapt zeros at various frequencies up to 3.5GHz with various filter characteristics. Thus, the integrated transversal filter can be used to mimic the inverse transfer function of dispersive communication channels for dispersion compensation. It can also be used a programmable notch filters in wireless transceiver designs.
I ,~~~l
0
0,5
1.5
2
2.5
FREQUENCY
3
3,5
4i5
4
id'
Fig. 10 Measured filter transfer function with bandrejection from 2GHz to 2.7GHz. The coefficients are set as 0, 60, 0, 25, 0, 100, 60 [mV]. The filter achieves a band-rejection of -20dB from 2GHz to 2.7GHz.
ACKNOWLEDGEMENTS
We acknowledge MOSIS for fabrication support under the MEP program. 45 26
-
6
i
lil
65
1
15
2
3 53
215
4
FREQUENCYx 0
f-
20
REFERENCES
45
-
[1] J. H. Winters, R. D. Gitlin and K. Sanjay, "Reducing the effects of Transmission Impairments in Digital Fiber Optic Systems," IEEE Communications Magazine, June 1993.
i-I--
Fig. 11 Measured filter transfer function withnochatd - - --.GHz. - - p sb- - - - -The -- - - m ag e - -- - setaslO,-in- --are 2eetinfom23 t- -p-- - -coefficients fle -- - -- -- - --- - - - - - -5-eh asOm.600.55m,0. 20m. The [magniTude of the nochievs l 1# 1.5 2 25 3 47b,a0 0.hchpovds-3d-nthrejection comparedom2GH to2.Gz 2.Gz Th fite coficet ar se as10m -0m 10m0 55m 0, 20m Th mantd fh oc is h jec__ 47d__ __wh__ic__h __pro_vi__des__ __33d no__ tc__re__ ti_on c_om__ re__ d o pa__ thasbn mgitudpe. __
__
__
__
__
Figure 8 demonstrates the filter characteristic with double notches at 2.3 GHz and 3.3 GHz. The magnitude of the notch is -55dB, which provides -37dB notch rejection compared to the pass band magnitude. Figure 9 shows the single notch characteristic of the tunable filter. Figure 10 illustrates the band-rejection characteristic of the filter, in which the filter achieves a band-rejection of -20dB from 2GHz to 2.7GHz. The magnitude of the notch is -43db, which provides -30db
____ __
[2] H. Wu, J. A. Tierno, P. Pepeljugoski, J. Schwab, S. Gowda, J.A. Kash, A. Hajimiri, "Integrated Transversal Equalizers in High-Speed Fiber-Optic Systems," IEEE J. Solid State Circuits, vol.38, pp. 213 1-2137, 2003.
[3] H. M. Rein and M. Moller, "Design Considerations for very high speed Si-bipolar IC's operating up to 50Gb/s", IEEE J. Solid State Circuits, vol.3 1, pp. 1076-1090, August 1996.
A 3.5GHz Low Power Programmable Transversal Filter RFIC Implemented in 47GHz SiGe Technology Vasanth Kakani, Xuefeng Yu, Foster F. Dai, Richard C. Jaeger
Department of Electrical and Computer Engineering Auburn University, Auburn, AL 36849-5201, USA
II.
Abstract This paper presents the design of a low power 3.5GHz analog programmable filter RFIC. The RF filter is a 7-tap transversal equalizer with cascaded Cherry-Hooper amplifiers for delay stages and Gilbert variable gain amplifier as tap weights. The delay stage using active devices greatly reduces the die area comparing to passive delay lines. The SiGe programmable filter RFIC consumes 250mW under 3.3V supply and occupies total 2.16mm2 die size.
Index Terms SiGe, transversal filter, analog filter, Cherry-Hooper amplifier.
TRANSVERSAL FILTER DESIGN
The block diagram of the implemented transversal filter is shown in Fig. 1. The RF filter includes an analog tapped delay line with the feed forward taps forming an FIR filter. The transfer function of the integrated filter can be adaptively adjusted by changing its tap weights. Changing the tap weights affects only the locations of the zero's, while the poles of the programmable filter are fixed. Hence the filter is always stable.
Filter Input
I. INTRODUCTION
Programmable RF filters find numerous applications in communication systems. In fiber communications, modal, chromatic and polarization mode dispersions are the major sources of transmission impairments. Electronic transversal filters can be used to compensate fiber dispersions by constructing an inverse transfer function of the dispersive channel [1]. For multi-band wireless transceiver designs, programmable RF notch filters are needed to selectively reject the bands based on various wireless standards. RF notch filters are critical for removing unwanted signals such as images and interferers. In [2], a fractionally spaced transversal filter is designed using passive transmission lines as delay elements. Passive delay elements occupy large die area and provide accurate delays only for a narrow frequency band. In this paper, we present a 3.5GHz programmable filter RFIC designed in 45GHz SiGe technology. In stead of using passive delay lines, we propose to use Cherry-Hooper amplifiers as the delay stages, which provide wide bandwidth and occupy small area. We used Gilbert variable gain amplifier for continuous tuning of the tap weights. Measured S21 of the programmable RF filter demonstrates various filter characteristics such low-pass, high pass and notch filters up to 3.5GHz frequency.
1-4244-0459-2/06/$20.00 ©2006 IEEE.
Filter Output Fig. 1. Block diagram of the transversal filter. T denotes the delay amplifier and ® denotes a variable gain amplifier.
Passive delay networks are either lossy (RC delay lines) or bulky (LC delay lines). Moreover, passive delay networks are always narrow band. In stead of using passive delay stages, we chose the series shunt cascaded Cherry-Hooper amplifier to implement the filter delay stages. As shown in Fig. 2, Cherry-Hooper amplifier [3] is a cascade of two feedback amplifiers, where the series feedback stage is a transconductance amplifier and shunt feedback stage is a trans-impedance amplifier. The amplifier with series feedback is driven from a low resistance voltage input obtained from the output of the shunt feedback stage. Conversely, the amplifier with shunt feedback is driven from a high resistance current input obtained from the output of the serial feedback stage. This arrangement is advantageous since the impedance requirement will be automatically
IEEE BCTM 4.4
satisfied at the input and output of the amplifier while cascading several such delay stages. Emitter followers are used in between the delay stages for level shifting and creating stronger impedance mismatch between succeeding stages to improve the bandwidth. Referred to Fig.2, the transfer function of the CherryHooper amplifier can be approximated as
k)O+SCF RI +g, -RF) Vi, 9,,3(1+S CF RF)(+S CE RE +9ml RE)
kOUI =g('+S CE
(1)
where RE and CE are the degeneration resistance and capacitance, respectively; RF and CF are the shunt feedback resistance and capacitance, respectively. The degeneration and feedback capacitors CE and CF introduce zeros in the frequency response and thereby maximize the amplifier bandwidth. The first zero of the amplifier frequency response is created by the degeneration capacitor CE, and the second zero is generated by the shunt feedback capacitor CF. Fig. 3 gives the simulated magnitude response of the amplifier. As shown, the 3-dB cutoff frequency of the amplifier is about 9 GHz.
Fig. 3 Simulated magnitude response of the delay stage with 3-dB bandwidth at 9 GHz
Figure 5 shows the small-signal model of the gain control circuit in the Gilbert cell. The Gilbert cell is a current amplifier with the current gain transfer function given as
I(S) __t9(1+SCIr.)-g,(1+SC,,3r,)l {glni r3+ss
g.4 (2)
RE
dc
RE
Fig. 4 Tap weights implemented using Gilbert variable gain amplifier. Fig. 2 Cherry--Hooper amplifier used to implement delay stages.
As shown in Fig. 4, the transversal filter tap with programmable gain is implemented using Gilbert variable gain amplifier. Tap weights are continuously adjustable between 0 and 1 of the CML logic level (about +100mV differential). Thus the variable gain stage is infact a variable loss stage. Flipping the polarity of the gain control signal Vage provides a phase shift of 1800 for negative tap coefficients.
Figure 6 shows the simulated magnitude response of the Gilbert variable gain amplifier. The 3-dB cutoff frequency is achieved at 14.5 GHz.
Finally the summation required in the transversal filter is performed in the current mode. The output current signals of all the taps are tied together to an external pull up resistor via a current buffer, which is a common-base amplifier.
IEEE BCTM 4.4
tati'at
constant current sources for the amplifiers. The filter RFIC also includes an input buffer, output buffers and CMOS to CML buffers. Figure 7 shows the die photo of the transversal filter RFIC.
The frequency response of the integrated filter was measured using a vector network analyzer. The measured filter transfer functions under different tap weights are shown in Fig. 8 to Fig. 11. lF~
-b3 l-
lin
-10
Fig. 5 Small-signal model of the gain control circuit.
--------------
-15 -26
0.0 I-, (NJ
U)
-1 .0
(D-
-30
-40
---
--
,15
-4,0 -50
0.00
5.00G
10AG
freq ( Hz )
Fig.6 Simulated magnitude response of the Gilbert variable gain amplifier. The 3-dB frequency is 14.5 GHz.
II
1
15
2
*-
2.5
45
4
3
FREQUENCY
x 100
Fig. 8 Measured filter transfer function with double notches at 2.3GHz and 3.3 GHz. The tap coefficients are set as -40, 75, -40, 75, -40, 75, 90 [mV]. The magnitude of the notch is -55dB, which provides -37dB notch rejection compared to the pass band magnitude.
IV. MEASURED RESULTS 10
--
-15
30~--
1
-- - --
f
The 3.5 GHz transversal filter is implemented in a 45GHz SiGe technology with a total 2.16mm2 die area including pads. The chip consumes 250mW power under a 3.3V supply voltage. The filter RFIC includes a bandgap reference to provide temperature independent
--E --
_
-n- -- -- - -- r- -- -- - -- -n-- ----- ir-- -- - -- -,--- -- - -- -T- -- -- - -- -1--- -- - -- -T-- -- -- ---
FREQUENCY Fig. 7. Die photo of the integrated transversal filter RFIC.
-----
x
loo
Fig. 9 Measured filter transfer function with notch at 2GHz. The filter coefficients are set as -85, 30, -20, 0, 30, 0, 0 [mV]. The magnitude of the notch is -43db, which provides -30db notch rejection compared to the pass band magnitude.
IEEE BCTM 4.4
notch rejection compared to the pass band magnitude. Thus, the integrated programmable RF filter is able to tune zeros to any frequency up to 3.5 GHz. V. CONCLUSIONS
We have implemented a low power 3.5GHz analog transversal filter in a 47GHz SiGe technology. The RF filter utilizes cascaded Cherry-Hooper amplifiers for delay stages and Gilbert variable gain amplifier as for continuous gain tuning. The delay stage using active devices greatly reduces the die area comparing to passive delay lines. Measured results show that by adjusting the tap coefficients, the integrated programmable filter IC is capable to adapt zeros at various frequencies up to 3.5GHz with various filter characteristics. Thus, the integrated transversal filter can be used to mimic the inverse transfer function of dispersive communication channels for dispersion compensation. It can also be used a programmable notch filters in wireless transceiver designs.
I ,~~~l
0
0,5
1.5
2
2.5
FREQUENCY
3
3,5
4i5
4
id'
Fig. 10 Measured filter transfer function with bandrejection from 2GHz to 2.7GHz. The coefficients are set as 0, 60, 0, 25, 0, 100, 60 [mV]. The filter achieves a band-rejection of -20dB from 2GHz to 2.7GHz.
ACKNOWLEDGEMENTS
We acknowledge MOSIS for fabrication support under the MEP program. 45 26
-
6
i
lil
65
1
15
2
3 53
215
4
FREQUENCYx 0
f-
20
REFERENCES
45
-
[1] J. H. Winters, R. D. Gitlin and K. Sanjay, "Reducing the effects of Transmission Impairments in Digital Fiber Optic Systems," IEEE Communications Magazine, June 1993.
i-I--
Fig. 11 Measured filter transfer function withnochatd - - --.GHz. - - p sb- - - - -The -- - - m ag e - -- - setaslO,-in- --are 2eetinfom23 t- -p-- - -coefficients fle -- - -- -- - --- - - - - - -5-eh asOm.600.55m,0. 20m. The [magniTude of the nochievs l 1# 1.5 2 25 3 47b,a0 0.hchpovds-3d-nthrejection comparedom2GH to2.Gz 2.Gz Th fite coficet ar se as10m -0m 10m0 55m 0, 20m Th mantd fh oc is h jec__ 47d__ __wh__ic__h __pro_vi__des__ __33d no__ tc__re__ ti_on c_om__ re__ d o pa__ thasbn mgitudpe. __
__
__
__
__
Figure 8 demonstrates the filter characteristic with double notches at 2.3 GHz and 3.3 GHz. The magnitude of the notch is -55dB, which provides -37dB notch rejection compared to the pass band magnitude. Figure 9 shows the single notch characteristic of the tunable filter. Figure 10 illustrates the band-rejection characteristic of the filter, in which the filter achieves a band-rejection of -20dB from 2GHz to 2.7GHz. The magnitude of the notch is -43db, which provides -30db
____ __
[2] H. Wu, J. A. Tierno, P. Pepeljugoski, J. Schwab, S. Gowda, J.A. Kash, A. Hajimiri, "Integrated Transversal Equalizers in High-Speed Fiber-Optic Systems," IEEE J. Solid State Circuits, vol.38, pp. 213 1-2137, 2003.
[3] H. M. Rein and M. Moller, "Design Considerations for very high speed Si-bipolar IC's operating up to 50Gb/s", IEEE J. Solid State Circuits, vol.3 1, pp. 1076-1090, August 1996.
