Microwave-Circuit-Embedded Resin Printed ... - Semantic Scholar
IEICE TRANS. ELECTRON., VOL.E88–C, NO.1 JANUARY 2005
83
PAPER
Special Section on Recent Trends of Microwave and Millimeter-Wave Passive Circuit Components
Microwave-Circuit-Embedded Resin Printed Circuit Board for Short Range Wireless Interfaces Akira SAITOU†a) , Kazuhiko HONJO†† , Kenichi SATO† , Toyoko KOYAMA† , and Koichi WATANABE† , Members
SUMMARY Microwave circuits embedded in a multi-layer resin PCB are demonstrated using low loss resin materials. Resin materials for microwave frequencies were compared with conventional FR-4 with respect to dielectric and conductor loss factors, which proved that losses could be reduced drastically with the low loss material and design optimizations. Baluns, switches and BPFs were designed and fabricated to estimate microwave performances. Measured and simulated insertion losses of the circuits for 2.5 GHz band, were 0.3 dB for a switch, 0.4 dB for a balun and 2.0 dB for a 3-stage Chebyshev BPF. An integration of a switch, a BPF and two baluns was successfully implemented in a multi-layer PCB. Insertion losses of the fabricated integrated circuit were less than 3 dB with 0.1 dB additional loss compared with a sum of individual circuit losses. With estimated results of temperature characteristics and reliability as well as low loss performances, microwave circuits in resin PCBs can be considered as a viable candidate for microwave equipments. key words: microwave, resin, PCB, balun, BPF, loss, integration
1.
Introduction
Recent remarkable popularization of mobile wireless communication equipments demands a low cost integration of microwave circuits in the smallest area. The integration of wireless circuits in a multi-layer resin PCB (Printed Circuit Board) is very advantageous. (1) Additional parts can be eliminated, and only a PCB with a semiconductor integrated circuit (IC) chip can constitute a RF block of equipments. (2) Unnecessary radiation can be drastically reduced because microwave circuits are confined between grounded conductors. (3) Low material cost (4) Performances also can be improved because connection losses with other components are reduced. However, because of extremely high loss tangent of conventional resin materials, few investigations have been implemented with resin materials [1], [2], and most of the conventional investigations for practical microwave circuits were implemented with low loss but more expensive Tefron or ceramic materials. Especially, LTCC (Low -Temperature Co-fired Ceramic) has been widely used for low loss microwave circuit integrations [3]–[5] to reduce the area. This paper demonstrates low loss performances of elementary microwave circuits of baluns, switches and bandpass filters (BPFs) with low loss resin PCB materials and relevant design optimizations. An integration of these cirManuscript received April 28, 2004. Manuscript revised August 9, 2004. † The authors are with YKC Corporation, Musashimurayamashi, 208-0023 Japan. †† The author is with the University of Electro-Communication, Chofu-shi, 182-8585 Japan. a) E-mail: [email protected]
cuits is also shown to be successfully implemented in a multi-layer resin PCB with little additional loss. The ambient temperature rise by IC’s power dissipation must be considered, because these circuits are assumed to be allocated under a semiconductor IC to minimize an area. Temperature characteristics and reliabilities of microwave circuits in the resin PCBs are shown to be very affirmative for practical microwave applications [6]. 2.
Loss Estimation of the Resin Materials
To obtain satisfactory performances in microwave frequencies, low loss tangent material is indispensable. Recently, resin materials for microwave frequencies and high speed transmission applications have been developed. To estimate the feasibility of low loss tangent resin materials which also accommodate conventional multi-layer fabrication processes, MegtronTM , Megtron5TM , FR-4, developed by Panasonic, were assessed. Strip transmission lines were fabricated in 422 µm-thick materials as shown in Fig. 1, and S-parameters were measured up to 20 GHz with an Agilent 8722ESTM network analyzer. To find the precise material parameters, measured S-parameters were compensated for measuring pad contributions. The compensated S-parameters were compared with the following analytical expressions of ideal transmission line S-parameters in order to find permittivities, loss tangents and conductor losses of the materials. S 11 = S 22 =
Z/Z0 − Z0 /Z Z/Z0 + Z0 /Z − 2 j cot(βL)
(1)
Fig. 1 Photograph and cross-section of the fabricated strip line. Two pads on both sides are used to measure S-parameters with Cascade onwafer probes. The inner strip line is 23.6 mm long and 0.1 mm wide. The resin thickness is 0.422 mm and the strip line of 22 µm-thick Cu,is in the middle of the resin.
c 2005 The Institute of Electronics, Information and Communication Engineers Copyright
IEICE TRANS. ELECTRON., VOL.E88–C, NO.1 JANUARY 2005
84
Fig. 3 terials.
Estimated relative permittivity vs frequency for different resin ma-
Fig. 2 Measure S-parameters (compensated for pad contributions) and ideal line analytical S-parameters.
Table 1
Strip line characteristics of resin materials at 2.5 GHz. Fig. 4 Comparison of the maximum available gain of the strip line for different resin materials.
2 S 21 = S 12 = j(Z/Z0 + Z0 /Z) sin(βL) + 2 cos(βL) β = 2π f εr (1 + j tan δ)/c + jαc f / f0
(2) (3)
Z0 ; port impedance (50 Ω) Z; characteristic impedance of a strip line β; complex phase constant L; transmission line length εr ; relative permittivity c; velocity of light in vacuum αc ; conductor loss coefficient f0 ;normalizing frequency (1 GHz in this paper) Fig. 2 shows the comparisons between ideal line Sparameters calculated with equations (1)–(3) and measured S-parameters compensated for pad contributions. The compensation of pad contributions was implemented by subtracting two pad T-parameters, found with electro- magnetic simulations, from the measured Tparameters. The material parameters were found by fitting ideal line S-parameters of varied parameters with the compensated measured S-parameters. Measured and compensated S-parameters can be fitted well with the ideal line analytical S-parameters as shown in Fig. 2. Table 1 summarizes the estimated parameter values of the materials found by fitting calibrated measured data and analytical expression of formulae (1)–(3) below 10 GHz. The loss tangent of Megtron5 was 0.004 and was reduced
to less than 1/4 compared with that of conventional FR4(0.018). Frequency dependence on the relative permittivity can be found with equation (2). After transforming the compensated S-parameters to the S-parameters terminated by transmission line characteristic impedance, relative permittivity can be found with the following formula. S 21 = S 12 =
1 = exp(− jβL) j sin(βL) + cos(βL)
(4)
Fig. 3 shows the estimated frequency dependences of the materials’ relative permittivities using equation (4). The relative permittivities of Megtron and Megtron5 were almost the same between 3.6 and 3.4 from 1 GHz to 10 GHz, and the relative permittivity of FR-4 was between 4.4 and 4.1. To compare the losses directly, maximum available gain of each fabricated strip line, which excludes return loss contributions and is a measure for net dielectric and conductor total loss, is shown in Fig. 4. Transmission line loss of Megtron5 is shown to be reduced clearly compared with that of conventional FR-4. 3.
Design and Fabrication of the Circuits
Filters, baluns and switches were designed and fabricated for 2.5 GHz, 5.2 GHz and 10 GHz bands using Megtron5 material. The thickness of the resin was selected to be 422 µm for all circuits. Fig. 5 shows the equivalent circuits of the designed balun and switch. For the Zin :Zout balun, impedance transform from Zin to Zout is accomplished with
SAITOU et al.: MICROWAVE-CIRCUIT-EMBEDDED RESIN PRINTED CIRCUIT BOARD FOR SHORT RANGE WIRELESS INTERFACES
85
Fig. 7 Measured and simulated characteristics of the switch On-state insertion loss was 0.3 dB.
O denotes open and S denotes short at each point. Diode’s state is either S or O and the impedance of each point changes according to each diode’s state. Fig. 5 Equivalent circuits of the designed balun and switch.
Fig. 6
Layouts of the circuits.
λ/4 transmission lines. The phase difference of π is accomplished with λ/2 transmission line as shown in Fig. 5(a). The single pole double throw (SPDT) switch is controlled to be on and off according to the diode’s on and off states, as shown in Fig. 5(b). The impedance is kept matched to 50 Ω along the on-state path. The BPFs were designed on 3-stage Chebyshev BPFs and were folded in hairpins to reduce the area [7]–[9]. To design these circuits, analytical transmission line dimensions were found first [10], [11], and dimensions were tuned with the ADSTM circuit simulator. Fig. 6 shows the layouts of the designed circuits. To
estimate the net loss of the switch, short state was realized by connecting to ground plane with an interstitial via instead of connecting to a diode. Measured and simulated characteristics of a fabricated switch, designed for 2.5 GHz ISM band, are shown in Fig. 7. One port was terminated by 50 Ω and 2-port S-parameters were measured with the other 2 ports. With three measurements by replacing 50 Ω terminated port, 3 × 3 S-parameters were obtained. On-state insertion loss was as small as 0.3 dB. Off-state attenuation and isolation between two output ports were less than −20 dB over the 2.5 GHz ISM band and the measured data corresponded well with the simulated data. Baluns of varied impedance transform ratios were designed and fabricated. The insertion losses were 0.4 dB for 1:1 and 1:2 baluns and 0.6 dB for 1:4 baluns which have another λ/4 50 Ω to 25 Ω impedance transformer in the input. With this transformer, transmission line impedances could be lowered to keep the minimum line width (80 µm) wide enough for mass production. The amplitude of S21 and S31 coincided at 2.3 GHz where insertion loss was less than 0.4 dB and phase difference became 180 degrees both for measured and simulated data as shown in Fig. 8(a). The return losses of the balanced port for a balanced input (Sdd22 ) and a common input (Scc22 ) can be obtained with 3 × 3 Sparameters terminated by Zin , Zout/2 , Zout/2 as follows. (5) Sdd22 = (S22 + S33 − S23 − S32 )/2 (6) Scc22 = (S22 + S33 + S23 + S32 )/2 Si j ; S-parameters terminated by Zin , Zout/2 , Zout/2 . Measured and simulated return losses for both unbalanced (S11 ) and balanced ports (Sdd22 ) were below −20 dB between 2.4 GHz and 2.5 GHz as shown in Fig. 8(b). Scc22 was −0.7 dB and unwanted common inputs were rejected by 85%. BPFs were designed and fabricated for 2.5 GHz, 5.2 GHz, and 10 GHz bands, and fractional bandwidths and ripples were varied. Fig. 9 shows measured and simulated characteristics of a BPF for which the fractional bandwidth was 10% and the ripple was 0.1 dB. Simulated and measured insertion losses and return losses agreed very well.
IEICE TRANS. ELECTRON., VOL.E88–C, NO.1 JANUARY 2005
86
Fig. 10 shows the trade-off relations between the insertion loss and the rejection estimated at a 18% lower frequency from the center frequency. BPFs for 10 GHz band exhibited a little bit lower insertion losses for the same rejection. For 30 dB off band rejection, insertion losses were just less than 4 dB which is considerably higher than the typical 2 dB loss for LTCC filters. To reduce an insertion loss, BPFs with thicker resin layers were designed and fabricated. The resin thickness was increased from 422 µm to 866 µm. The results are also shown in Fig. 10. With the same 30 dB off band rejection, the insertion loss was reduced to 2.0 dB due to dielectric loss reduction because of the increased resin substrate thickness and due to conductor loss reduction because of the 2.1 times wider lines’ reduced resistances. The breakdown of the losses was quantified by simulations to be 0.6 dB dielectric loss, 0.4 dB additional conductor loss by surface roughness and 1.0 dB conductor loss of copper’s 5.8×107 finite conductivity, which showed that the total loss could be kept low with design optimizations in spite of a little bit higher loss tangent of a resin material compared with ceramics. 4. Fig. 8
Measured and simulated return losses of 1: 2 balun.
Integration of the Circuits
A BPF, a switch and two baluns were integrated in a multilayer PCB as shown in Fig. 11, Fig. 12 and Fig. 13. A BPF was allocated in an 866 µm thick layer to reduce an insertion loss. Two baluns and a switch were allocated in upper and lower layers separately to reduce an area. A 1:1 balun was placed in the upper layer and a 1:4 balun was placed in the lower layer. Connections between layers were implemented with interstitial via holes which only penetrated intermediate layers between two connected layers. Total size of the circuits was 18 mm × 12 mm × 0.87 mm as shown in Fig. 13.
Fig. 9 Measured and simulated characteristics of 3stage Chebyshev BPF. The fractional bandwidth is 10% and the ripple is 0.1 dB.
Fig. 11
Fig. 10 Trade-off between an insertion loss and a rejection. BPFs were designed for different bands, fractional bandwidths, and ripples. Resin substrate thickness of the BPFs is shown in the figure. The rejections were estimated at 2.0 GHz, 4.3 GHz, and 8.2 GHz for 2.45 GHz, 5.2 GHz and 10 GHz bands.
Fig. 12
Configuration of the integrated circuit.
Layout and cross-section of the integrated circuits.
SAITOU et al.: MICROWAVE-CIRCUIT-EMBEDDED RESIN PRINTED CIRCUIT BOARD FOR SHORT RANGE WIRELESS INTERFACES
87
Fig. 13 Photograph of the integrated circuits. The area of a integrated PCB is 18 mm × 12 mm × 0.87 mm. Fig. 15
Fig. 16 BPF.
Fig. 14 circuit.
Temperature characteristics of insertion loss and rejection of the
Measured and simulated characteristics of on-state integrated
Measured and simulated microwave performances of the integrated circuit are shown in Fig. 14. The total insertion loss of a BPF, an on-state switch and a balun was 2.8 dB for the 1:1 balun channel and 3.0 dB for the 1:4 balun channel, which are almost the same with the summed insertion losses of each circuit (2.7 dB and 2.9 dB). The return losses of the integrated circuit was less than −10 dB for both balanced and unbalanced ports over the 2.5 GHz ISM band. 5.
Temperature characteristics of S21 of the BPF.
Temperature Characteristics and Reliability
To investigate effects of the temperature rise by IC’s power dissipation, temperature characteristics of microwave performances in resin PCBs were estimated for BPFs of 2.5 GHz band between 23◦ C and 200◦ C. Temperature characteristics of insertion losses and rejections are shown in Fig. 15 and Fig. 16. Insertion loss at 2.45 GHz increased by 0.36 dB from 23◦ C to 200◦ C (2.0 × 10−3 dB/deg). Rejections at 2 GHz and 2.9 GHz changed less than 0.5 dB as shown in Fig. 16. The center frequency shifted a little bit lower up to 160◦ C as shown in fig.15, and the lower 2.0 GHz
Fig. 17 Insertion loss and attenuation shift of the BPFs by 150◦ C high temperature storage. Three BPFs were designed for 2.5 GHz (), 5.2 GHz () and 10 GHz (♦) bands. Ripples and fractional bandwidths were designed to be 0.1 dB and 10% in each band.
attenuation reduced in spite of the loss increase by higher temperature. The decrease of the center frequency was attributed by the thermal expansion of the resin of which catalogue data was 60 ppm/deg. Line length expansion caused by 133deg temperature rise was 0.8%, which explained well the band center frequency decrease of 1%. Therefore, the permittivity was almost the same below 160◦ C. However, above 160◦ C the band center frequency increased, and the permittivity was estimated to decrease above 160◦ C, which also explained the reduction of 2.9 GHz attenuation beyond 160◦ C. High temperature storage at 150◦ C was implemented to investigate microwave performance degradations. Fig. 17 shows the shifts of insertion losses and rejections, and no
IEICE TRANS. ELECTRON., VOL.E88–C, NO.1 JANUARY 2005
88
systematic changes were observed before 1000 hours. These results demonstrate that microwave circuits integrated in the low loss resin PCBs can serve practical use in wireless equipments. 6.
Conclusion
Microwave circuits embedded in a multi-layer resin PCB are demonstrated using low loss resin materials. Low loss resin materials were compared with conventional FR-4 with respect to dielectric and conductor loss factors, which proved that losses could be reduced drastically with the low loss material and design optimizations. Baluns, switches and BPFs were designed and fabricated to estimate microwave performances. Both of measured and simulated insertion losses of the circuits for 2.5 GHz band, were 0.3 dB for a switch, 0.4 dB for a balun and 2.0 dB for a 3-stage Chebyshev BPF. Simulated and measured data agreed well, which showed the design accuracy of microwave circuits in a lowloss resin material. An integration of a switch, a BPF and two baluns was successfully implemented in a multi-layer PCB. The insertion losses of the fabricated integrated circuit were less than 3 dB with 0.1 dB additional loss compared with a sum of individual circuit losses. With estimated results of temperature characteristics and reliability as well as low loss performances, microwave circuits in multi-layer resin PCBs can be considered as a viable candidate for microwave equipments.
Circuits, Artech House, London, 1999. [11] J.A.G. Malherbe, Microwave Transmission Line Couplers, Artech House, London, 1988. Akira Saitou received the B.E. and M.E. degrees in applied physics from the University of Tokyo, Tokyo, Japan, in 1975 and 1977 respectively. During 1977–2002, he stayed in NEC Corporation to develop GaAs FET’s and MMIC’s for microwave and millimeter wave communication. He presently is with YKC Corporation to develop microwave circuits and antennas for short range wireless interfaces. He is a member of the IEEE. Kazuhiko Honjo received the B.E. degree from the University of Electro- Communications, Tokyo, Japan, in 1974 and M.E. and D.E. degrees in electronic engineering from the Tokyo Institute of Technology, Tokyo, Japan, in 1976 and 1983, respectively. In 1976, he joined NEC corporation, Kawasaki, Japan. He has been involved in research and development of high power GaAs FET amplifiers, GaAs MMIC’s, HBT’s and their circuit applications for microwave and millimeter wave systems and fiber optic communication systems. In 2001 he moved to the University of Electro-Communications. He is now Professor of the university at the Department of Information and Communication Engineering. Prof. Honjo received both the 1983 Microwave Prize and 1988 Microwave Prize granted by IEEE MTT-S, the Young Engineer Award (1980), and the Society Award (Electronics Award, 1999) both from IEICE. He is Fellow of IEEE.
References [1] N. Imai, K. Honjo, and A. Saitou, “Basic consideration for variable bandwidth filters using resin printed circuit board,” Proc. IEICE Gen. Conf. 2003, C-2-87, 2003. [2] A. Saitou, K. Sato, T. Koyama, K. Watanabe, and K. Honjo, “L/S, C and X band passive- circuit-embedded resin printed circuit board,” Proc. IEICE Gen. Conf. 2003, C-2-67, 2003. [3] C. Tang and C. Chang, “A semi-lumped balun fabricated by low temperature co-fired ceramic,” IEEE MTT-S International Microwave Symposium Digest, pp.2201–2204, 2002. [4] C.H. Lee, S. Chakraborty, A. Sutono, S. Yoo, D. Heo, and J. Laskar, “Broadband highly integrated LTCC front-end moule for IEEE 802.11a WLAN applications,” IEEE MTT-S International Microwave Symposium Digest, pp.1045–1048, 2002. [5] T. Ohwada, H. Ikematu, H. Oh-hashi, T. Takagi, and O. Isida, “A Kuband low-loss stripline low-pass filter for LTCC modules with lowimpedance lines to obtain plural transmission zeros,” IEEE MTT-S International Microwave Symposium Digest, pp.1617–1620, 2002. [6] O. Kotaki, A. Saitou, K. Sato, T. Koyama, K. Watanabe, and K. Honjo “Temperature characteristics and reliability for microwave passive-circuit-embedded resin printed circuit board,” Proc. Electron. Conf. IEICE 2003, C-2-56, 2003. [7] S.B. Cohn, “Parallel-coupled transmission-line-resonator filters,” IRE Trans. Microw. Theory Tech., vol.MTT-6, no.2, pp.223–231, 1958. [8] E.G. Cristal and S. Frankel, “Hairpin-line and hybrid hairpinlike/half-wave parallel-coupled-line filters,” IEEE Trans. Microw. Theory Tech., vol.20, no.11, pp.719–728, 1972. [9] G. Matthaei, L. Young, and E.M.T. Jones, Microwave Filters, Impedance-Matching Networks, and Coupling Structures, Artech House, 1980. [10] R. Mongia, I. Bahl, and P. Bhartia, RF and Microwave Coupled-Line
Kenichi Sato graduated Information Processing Department of Nippon Engineering College of Hachioji, Tokyo Japan, in 1997. He joined YKC Corporation, Tokyo Japan, in1997 and has been involved in design, process development and applied technology of printed circuit boards.
Toyoko Koyama received the B.E. degree in techno-chemistry from the College of Industrial Technology, Nihon University, Chiba Japan, in 2000. She joined YKC Corporation, Tokyo Japan, in 2000 and has been involved in process development of printed circuit boards.
Koichi Watanabe received the B.E. degree in techno-chemistry from the College of Science and Technology, Nihon University, Tokyo Japan, in 1976. He joined YKC Corporation, Tokyo Japan, in 1976 and has been involved in process development and reliability engineering. He is currently a general manager of Technology Department and Reliability Department.
83
PAPER
Special Section on Recent Trends of Microwave and Millimeter-Wave Passive Circuit Components
Microwave-Circuit-Embedded Resin Printed Circuit Board for Short Range Wireless Interfaces Akira SAITOU†a) , Kazuhiko HONJO†† , Kenichi SATO† , Toyoko KOYAMA† , and Koichi WATANABE† , Members
SUMMARY Microwave circuits embedded in a multi-layer resin PCB are demonstrated using low loss resin materials. Resin materials for microwave frequencies were compared with conventional FR-4 with respect to dielectric and conductor loss factors, which proved that losses could be reduced drastically with the low loss material and design optimizations. Baluns, switches and BPFs were designed and fabricated to estimate microwave performances. Measured and simulated insertion losses of the circuits for 2.5 GHz band, were 0.3 dB for a switch, 0.4 dB for a balun and 2.0 dB for a 3-stage Chebyshev BPF. An integration of a switch, a BPF and two baluns was successfully implemented in a multi-layer PCB. Insertion losses of the fabricated integrated circuit were less than 3 dB with 0.1 dB additional loss compared with a sum of individual circuit losses. With estimated results of temperature characteristics and reliability as well as low loss performances, microwave circuits in resin PCBs can be considered as a viable candidate for microwave equipments. key words: microwave, resin, PCB, balun, BPF, loss, integration
1.
Introduction
Recent remarkable popularization of mobile wireless communication equipments demands a low cost integration of microwave circuits in the smallest area. The integration of wireless circuits in a multi-layer resin PCB (Printed Circuit Board) is very advantageous. (1) Additional parts can be eliminated, and only a PCB with a semiconductor integrated circuit (IC) chip can constitute a RF block of equipments. (2) Unnecessary radiation can be drastically reduced because microwave circuits are confined between grounded conductors. (3) Low material cost (4) Performances also can be improved because connection losses with other components are reduced. However, because of extremely high loss tangent of conventional resin materials, few investigations have been implemented with resin materials [1], [2], and most of the conventional investigations for practical microwave circuits were implemented with low loss but more expensive Tefron or ceramic materials. Especially, LTCC (Low -Temperature Co-fired Ceramic) has been widely used for low loss microwave circuit integrations [3]–[5] to reduce the area. This paper demonstrates low loss performances of elementary microwave circuits of baluns, switches and bandpass filters (BPFs) with low loss resin PCB materials and relevant design optimizations. An integration of these cirManuscript received April 28, 2004. Manuscript revised August 9, 2004. † The authors are with YKC Corporation, Musashimurayamashi, 208-0023 Japan. †† The author is with the University of Electro-Communication, Chofu-shi, 182-8585 Japan. a) E-mail: [email protected]
cuits is also shown to be successfully implemented in a multi-layer resin PCB with little additional loss. The ambient temperature rise by IC’s power dissipation must be considered, because these circuits are assumed to be allocated under a semiconductor IC to minimize an area. Temperature characteristics and reliabilities of microwave circuits in the resin PCBs are shown to be very affirmative for practical microwave applications [6]. 2.
Loss Estimation of the Resin Materials
To obtain satisfactory performances in microwave frequencies, low loss tangent material is indispensable. Recently, resin materials for microwave frequencies and high speed transmission applications have been developed. To estimate the feasibility of low loss tangent resin materials which also accommodate conventional multi-layer fabrication processes, MegtronTM , Megtron5TM , FR-4, developed by Panasonic, were assessed. Strip transmission lines were fabricated in 422 µm-thick materials as shown in Fig. 1, and S-parameters were measured up to 20 GHz with an Agilent 8722ESTM network analyzer. To find the precise material parameters, measured S-parameters were compensated for measuring pad contributions. The compensated S-parameters were compared with the following analytical expressions of ideal transmission line S-parameters in order to find permittivities, loss tangents and conductor losses of the materials. S 11 = S 22 =
Z/Z0 − Z0 /Z Z/Z0 + Z0 /Z − 2 j cot(βL)
(1)
Fig. 1 Photograph and cross-section of the fabricated strip line. Two pads on both sides are used to measure S-parameters with Cascade onwafer probes. The inner strip line is 23.6 mm long and 0.1 mm wide. The resin thickness is 0.422 mm and the strip line of 22 µm-thick Cu,is in the middle of the resin.
c 2005 The Institute of Electronics, Information and Communication Engineers Copyright
IEICE TRANS. ELECTRON., VOL.E88–C, NO.1 JANUARY 2005
84
Fig. 3 terials.
Estimated relative permittivity vs frequency for different resin ma-
Fig. 2 Measure S-parameters (compensated for pad contributions) and ideal line analytical S-parameters.
Table 1
Strip line characteristics of resin materials at 2.5 GHz. Fig. 4 Comparison of the maximum available gain of the strip line for different resin materials.
2 S 21 = S 12 = j(Z/Z0 + Z0 /Z) sin(βL) + 2 cos(βL) β = 2π f εr (1 + j tan δ)/c + jαc f / f0
(2) (3)
Z0 ; port impedance (50 Ω) Z; characteristic impedance of a strip line β; complex phase constant L; transmission line length εr ; relative permittivity c; velocity of light in vacuum αc ; conductor loss coefficient f0 ;normalizing frequency (1 GHz in this paper) Fig. 2 shows the comparisons between ideal line Sparameters calculated with equations (1)–(3) and measured S-parameters compensated for pad contributions. The compensation of pad contributions was implemented by subtracting two pad T-parameters, found with electro- magnetic simulations, from the measured Tparameters. The material parameters were found by fitting ideal line S-parameters of varied parameters with the compensated measured S-parameters. Measured and compensated S-parameters can be fitted well with the ideal line analytical S-parameters as shown in Fig. 2. Table 1 summarizes the estimated parameter values of the materials found by fitting calibrated measured data and analytical expression of formulae (1)–(3) below 10 GHz. The loss tangent of Megtron5 was 0.004 and was reduced
to less than 1/4 compared with that of conventional FR4(0.018). Frequency dependence on the relative permittivity can be found with equation (2). After transforming the compensated S-parameters to the S-parameters terminated by transmission line characteristic impedance, relative permittivity can be found with the following formula. S 21 = S 12 =
1 = exp(− jβL) j sin(βL) + cos(βL)
(4)
Fig. 3 shows the estimated frequency dependences of the materials’ relative permittivities using equation (4). The relative permittivities of Megtron and Megtron5 were almost the same between 3.6 and 3.4 from 1 GHz to 10 GHz, and the relative permittivity of FR-4 was between 4.4 and 4.1. To compare the losses directly, maximum available gain of each fabricated strip line, which excludes return loss contributions and is a measure for net dielectric and conductor total loss, is shown in Fig. 4. Transmission line loss of Megtron5 is shown to be reduced clearly compared with that of conventional FR-4. 3.
Design and Fabrication of the Circuits
Filters, baluns and switches were designed and fabricated for 2.5 GHz, 5.2 GHz and 10 GHz bands using Megtron5 material. The thickness of the resin was selected to be 422 µm for all circuits. Fig. 5 shows the equivalent circuits of the designed balun and switch. For the Zin :Zout balun, impedance transform from Zin to Zout is accomplished with
SAITOU et al.: MICROWAVE-CIRCUIT-EMBEDDED RESIN PRINTED CIRCUIT BOARD FOR SHORT RANGE WIRELESS INTERFACES
85
Fig. 7 Measured and simulated characteristics of the switch On-state insertion loss was 0.3 dB.
O denotes open and S denotes short at each point. Diode’s state is either S or O and the impedance of each point changes according to each diode’s state. Fig. 5 Equivalent circuits of the designed balun and switch.
Fig. 6
Layouts of the circuits.
λ/4 transmission lines. The phase difference of π is accomplished with λ/2 transmission line as shown in Fig. 5(a). The single pole double throw (SPDT) switch is controlled to be on and off according to the diode’s on and off states, as shown in Fig. 5(b). The impedance is kept matched to 50 Ω along the on-state path. The BPFs were designed on 3-stage Chebyshev BPFs and were folded in hairpins to reduce the area [7]–[9]. To design these circuits, analytical transmission line dimensions were found first [10], [11], and dimensions were tuned with the ADSTM circuit simulator. Fig. 6 shows the layouts of the designed circuits. To
estimate the net loss of the switch, short state was realized by connecting to ground plane with an interstitial via instead of connecting to a diode. Measured and simulated characteristics of a fabricated switch, designed for 2.5 GHz ISM band, are shown in Fig. 7. One port was terminated by 50 Ω and 2-port S-parameters were measured with the other 2 ports. With three measurements by replacing 50 Ω terminated port, 3 × 3 S-parameters were obtained. On-state insertion loss was as small as 0.3 dB. Off-state attenuation and isolation between two output ports were less than −20 dB over the 2.5 GHz ISM band and the measured data corresponded well with the simulated data. Baluns of varied impedance transform ratios were designed and fabricated. The insertion losses were 0.4 dB for 1:1 and 1:2 baluns and 0.6 dB for 1:4 baluns which have another λ/4 50 Ω to 25 Ω impedance transformer in the input. With this transformer, transmission line impedances could be lowered to keep the minimum line width (80 µm) wide enough for mass production. The amplitude of S21 and S31 coincided at 2.3 GHz where insertion loss was less than 0.4 dB and phase difference became 180 degrees both for measured and simulated data as shown in Fig. 8(a). The return losses of the balanced port for a balanced input (Sdd22 ) and a common input (Scc22 ) can be obtained with 3 × 3 Sparameters terminated by Zin , Zout/2 , Zout/2 as follows. (5) Sdd22 = (S22 + S33 − S23 − S32 )/2 (6) Scc22 = (S22 + S33 + S23 + S32 )/2 Si j ; S-parameters terminated by Zin , Zout/2 , Zout/2 . Measured and simulated return losses for both unbalanced (S11 ) and balanced ports (Sdd22 ) were below −20 dB between 2.4 GHz and 2.5 GHz as shown in Fig. 8(b). Scc22 was −0.7 dB and unwanted common inputs were rejected by 85%. BPFs were designed and fabricated for 2.5 GHz, 5.2 GHz, and 10 GHz bands, and fractional bandwidths and ripples were varied. Fig. 9 shows measured and simulated characteristics of a BPF for which the fractional bandwidth was 10% and the ripple was 0.1 dB. Simulated and measured insertion losses and return losses agreed very well.
IEICE TRANS. ELECTRON., VOL.E88–C, NO.1 JANUARY 2005
86
Fig. 10 shows the trade-off relations between the insertion loss and the rejection estimated at a 18% lower frequency from the center frequency. BPFs for 10 GHz band exhibited a little bit lower insertion losses for the same rejection. For 30 dB off band rejection, insertion losses were just less than 4 dB which is considerably higher than the typical 2 dB loss for LTCC filters. To reduce an insertion loss, BPFs with thicker resin layers were designed and fabricated. The resin thickness was increased from 422 µm to 866 µm. The results are also shown in Fig. 10. With the same 30 dB off band rejection, the insertion loss was reduced to 2.0 dB due to dielectric loss reduction because of the increased resin substrate thickness and due to conductor loss reduction because of the 2.1 times wider lines’ reduced resistances. The breakdown of the losses was quantified by simulations to be 0.6 dB dielectric loss, 0.4 dB additional conductor loss by surface roughness and 1.0 dB conductor loss of copper’s 5.8×107 finite conductivity, which showed that the total loss could be kept low with design optimizations in spite of a little bit higher loss tangent of a resin material compared with ceramics. 4. Fig. 8
Measured and simulated return losses of 1: 2 balun.
Integration of the Circuits
A BPF, a switch and two baluns were integrated in a multilayer PCB as shown in Fig. 11, Fig. 12 and Fig. 13. A BPF was allocated in an 866 µm thick layer to reduce an insertion loss. Two baluns and a switch were allocated in upper and lower layers separately to reduce an area. A 1:1 balun was placed in the upper layer and a 1:4 balun was placed in the lower layer. Connections between layers were implemented with interstitial via holes which only penetrated intermediate layers between two connected layers. Total size of the circuits was 18 mm × 12 mm × 0.87 mm as shown in Fig. 13.
Fig. 9 Measured and simulated characteristics of 3stage Chebyshev BPF. The fractional bandwidth is 10% and the ripple is 0.1 dB.
Fig. 11
Fig. 10 Trade-off between an insertion loss and a rejection. BPFs were designed for different bands, fractional bandwidths, and ripples. Resin substrate thickness of the BPFs is shown in the figure. The rejections were estimated at 2.0 GHz, 4.3 GHz, and 8.2 GHz for 2.45 GHz, 5.2 GHz and 10 GHz bands.
Fig. 12
Configuration of the integrated circuit.
Layout and cross-section of the integrated circuits.
SAITOU et al.: MICROWAVE-CIRCUIT-EMBEDDED RESIN PRINTED CIRCUIT BOARD FOR SHORT RANGE WIRELESS INTERFACES
87
Fig. 13 Photograph of the integrated circuits. The area of a integrated PCB is 18 mm × 12 mm × 0.87 mm. Fig. 15
Fig. 16 BPF.
Fig. 14 circuit.
Temperature characteristics of insertion loss and rejection of the
Measured and simulated characteristics of on-state integrated
Measured and simulated microwave performances of the integrated circuit are shown in Fig. 14. The total insertion loss of a BPF, an on-state switch and a balun was 2.8 dB for the 1:1 balun channel and 3.0 dB for the 1:4 balun channel, which are almost the same with the summed insertion losses of each circuit (2.7 dB and 2.9 dB). The return losses of the integrated circuit was less than −10 dB for both balanced and unbalanced ports over the 2.5 GHz ISM band. 5.
Temperature characteristics of S21 of the BPF.
Temperature Characteristics and Reliability
To investigate effects of the temperature rise by IC’s power dissipation, temperature characteristics of microwave performances in resin PCBs were estimated for BPFs of 2.5 GHz band between 23◦ C and 200◦ C. Temperature characteristics of insertion losses and rejections are shown in Fig. 15 and Fig. 16. Insertion loss at 2.45 GHz increased by 0.36 dB from 23◦ C to 200◦ C (2.0 × 10−3 dB/deg). Rejections at 2 GHz and 2.9 GHz changed less than 0.5 dB as shown in Fig. 16. The center frequency shifted a little bit lower up to 160◦ C as shown in fig.15, and the lower 2.0 GHz
Fig. 17 Insertion loss and attenuation shift of the BPFs by 150◦ C high temperature storage. Three BPFs were designed for 2.5 GHz (), 5.2 GHz () and 10 GHz (♦) bands. Ripples and fractional bandwidths were designed to be 0.1 dB and 10% in each band.
attenuation reduced in spite of the loss increase by higher temperature. The decrease of the center frequency was attributed by the thermal expansion of the resin of which catalogue data was 60 ppm/deg. Line length expansion caused by 133deg temperature rise was 0.8%, which explained well the band center frequency decrease of 1%. Therefore, the permittivity was almost the same below 160◦ C. However, above 160◦ C the band center frequency increased, and the permittivity was estimated to decrease above 160◦ C, which also explained the reduction of 2.9 GHz attenuation beyond 160◦ C. High temperature storage at 150◦ C was implemented to investigate microwave performance degradations. Fig. 17 shows the shifts of insertion losses and rejections, and no
IEICE TRANS. ELECTRON., VOL.E88–C, NO.1 JANUARY 2005
88
systematic changes were observed before 1000 hours. These results demonstrate that microwave circuits integrated in the low loss resin PCBs can serve practical use in wireless equipments. 6.
Conclusion
Microwave circuits embedded in a multi-layer resin PCB are demonstrated using low loss resin materials. Low loss resin materials were compared with conventional FR-4 with respect to dielectric and conductor loss factors, which proved that losses could be reduced drastically with the low loss material and design optimizations. Baluns, switches and BPFs were designed and fabricated to estimate microwave performances. Both of measured and simulated insertion losses of the circuits for 2.5 GHz band, were 0.3 dB for a switch, 0.4 dB for a balun and 2.0 dB for a 3-stage Chebyshev BPF. Simulated and measured data agreed well, which showed the design accuracy of microwave circuits in a lowloss resin material. An integration of a switch, a BPF and two baluns was successfully implemented in a multi-layer PCB. The insertion losses of the fabricated integrated circuit were less than 3 dB with 0.1 dB additional loss compared with a sum of individual circuit losses. With estimated results of temperature characteristics and reliability as well as low loss performances, microwave circuits in multi-layer resin PCBs can be considered as a viable candidate for microwave equipments.
Circuits, Artech House, London, 1999. [11] J.A.G. Malherbe, Microwave Transmission Line Couplers, Artech House, London, 1988. Akira Saitou received the B.E. and M.E. degrees in applied physics from the University of Tokyo, Tokyo, Japan, in 1975 and 1977 respectively. During 1977–2002, he stayed in NEC Corporation to develop GaAs FET’s and MMIC’s for microwave and millimeter wave communication. He presently is with YKC Corporation to develop microwave circuits and antennas for short range wireless interfaces. He is a member of the IEEE. Kazuhiko Honjo received the B.E. degree from the University of Electro- Communications, Tokyo, Japan, in 1974 and M.E. and D.E. degrees in electronic engineering from the Tokyo Institute of Technology, Tokyo, Japan, in 1976 and 1983, respectively. In 1976, he joined NEC corporation, Kawasaki, Japan. He has been involved in research and development of high power GaAs FET amplifiers, GaAs MMIC’s, HBT’s and their circuit applications for microwave and millimeter wave systems and fiber optic communication systems. In 2001 he moved to the University of Electro-Communications. He is now Professor of the university at the Department of Information and Communication Engineering. Prof. Honjo received both the 1983 Microwave Prize and 1988 Microwave Prize granted by IEEE MTT-S, the Young Engineer Award (1980), and the Society Award (Electronics Award, 1999) both from IEICE. He is Fellow of IEEE.
References [1] N. Imai, K. Honjo, and A. Saitou, “Basic consideration for variable bandwidth filters using resin printed circuit board,” Proc. IEICE Gen. Conf. 2003, C-2-87, 2003. [2] A. Saitou, K. Sato, T. Koyama, K. Watanabe, and K. Honjo, “L/S, C and X band passive- circuit-embedded resin printed circuit board,” Proc. IEICE Gen. Conf. 2003, C-2-67, 2003. [3] C. Tang and C. Chang, “A semi-lumped balun fabricated by low temperature co-fired ceramic,” IEEE MTT-S International Microwave Symposium Digest, pp.2201–2204, 2002. [4] C.H. Lee, S. Chakraborty, A. Sutono, S. Yoo, D. Heo, and J. Laskar, “Broadband highly integrated LTCC front-end moule for IEEE 802.11a WLAN applications,” IEEE MTT-S International Microwave Symposium Digest, pp.1045–1048, 2002. [5] T. Ohwada, H. Ikematu, H. Oh-hashi, T. Takagi, and O. Isida, “A Kuband low-loss stripline low-pass filter for LTCC modules with lowimpedance lines to obtain plural transmission zeros,” IEEE MTT-S International Microwave Symposium Digest, pp.1617–1620, 2002. [6] O. Kotaki, A. Saitou, K. Sato, T. Koyama, K. Watanabe, and K. Honjo “Temperature characteristics and reliability for microwave passive-circuit-embedded resin printed circuit board,” Proc. Electron. Conf. IEICE 2003, C-2-56, 2003. [7] S.B. Cohn, “Parallel-coupled transmission-line-resonator filters,” IRE Trans. Microw. Theory Tech., vol.MTT-6, no.2, pp.223–231, 1958. [8] E.G. Cristal and S. Frankel, “Hairpin-line and hybrid hairpinlike/half-wave parallel-coupled-line filters,” IEEE Trans. Microw. Theory Tech., vol.20, no.11, pp.719–728, 1972. [9] G. Matthaei, L. Young, and E.M.T. Jones, Microwave Filters, Impedance-Matching Networks, and Coupling Structures, Artech House, 1980. [10] R. Mongia, I. Bahl, and P. Bhartia, RF and Microwave Coupled-Line
Kenichi Sato graduated Information Processing Department of Nippon Engineering College of Hachioji, Tokyo Japan, in 1997. He joined YKC Corporation, Tokyo Japan, in1997 and has been involved in design, process development and applied technology of printed circuit boards.
Toyoko Koyama received the B.E. degree in techno-chemistry from the College of Industrial Technology, Nihon University, Chiba Japan, in 2000. She joined YKC Corporation, Tokyo Japan, in 2000 and has been involved in process development of printed circuit boards.
Koichi Watanabe received the B.E. degree in techno-chemistry from the College of Science and Technology, Nihon University, Tokyo Japan, in 1976. He joined YKC Corporation, Tokyo Japan, in 1976 and has been involved in process development and reliability engineering. He is currently a general manager of Technology Department and Reliability Department.
